Circuit for Object Detection and Vehicle Position Determination

ABSTRACT

A multi-purpose detection circuit for object detection and vehicle position determination is described. For example, the circuit is configurable for detecting foreign metallic objects, living objects, and a vehicle or type of vehicle above an inductive wireless power transmitter. The circuit is also configurable for determining the vehicle&#39;s position relative to the inductive wireless power transmitter. An example apparatus includes a measurement circuit including a multiplexer, electrically connected to a plurality of inductive and capacitive sense circuits, for measuring one or more electrical characteristics in each of the inductive and capacitive sense circuits according to a predetermined time multiplexing scheme. The apparatus further includes a control and evaluation circuit for evaluating the measured electrical characteristics and determining at least one of a presence of a metallic object, a living object, a vehicle, or a type of vehicle, and a vehicle position based on changes in the measured electrical characteristics.

FIELD

The present disclosure relates generally to object detection and vehicleposition determination, for example, in an application for inductivewireless charging of electric vehicles. In particular, the presentdisclosure is directed to a multi-purpose detection circuit configurablefor detecting foreign metallic objects, living objects located near aninductive wireless power transmitter as well as for detecting a vehicleabove the wireless power transmitter and for determining a position ofthe vehicle relative to the inductive wireless power transmitter.

BACKGROUND

Inductive wireless power transfer (WPT) systems provide one example ofwireless transfer of energy. In an inductive WPT system, a primary powerdevice (or wireless power transmitter) transmits power wirelessly to asecondary power device (or wireless power receiver). Each of thewireless power transmitter and wireless power receiver includes aminductive power transfer structure, typically a single or multi-coilarrangement of windings comprising electric current conveying materials(e.g., copper Litz wire). An alternating current passing through thecoil e.g., of a primary wireless power transfer structure produces analternating magnetic field. When a secondary wireless power transferstructure is placed in proximity to the primary wireless power transferstructure, the alternating magnetic field induces an electromotive force(EMF) into the secondary wireless power transfer structure according toFaraday's law, thereby wirelessly transferring power to the wirelesspower receiver if a resistive load is connected to the wireless powerreceiver. To improve a power transfer efficiency, some implementationsuse a wireless power transfer structure that is part of a resonantstructure (resonator). The resonant structure may comprise acapacitively loaded inductor forming a resonance substantially at afundamental operating frequency of the inductive WPT system (e.g., inthe range from 80 kHz to 90 kHz).

Inductive wireless power transfer to electrically chargeable vehicles atpower levels of several kilowatts in both domestic and public parkingzones may require special protective measures for safety of persons andequipment. Such measures may include detection of foreign objects in aninductive power region of the inductive WPT system where electromagneticfield exposure levels exceed certain limits. This may be particularlytrue for systems where the inductive power region is open andaccessible. Such measures may include detection of electricallyconducting (metallic) objects and living objects, (e.g., humans,extremities of humans, or animals) that may be present within or nearthe inductive power region.

In certain applications for inductive wireless charging of electricvehicles, it may be useful to be able to detect foreign objects that maybe present in the inductive power region and that could be susceptibleto induction heating due to the high magnetic field strength in thatregion. In an inductive wireless power transfer system for electricvehicle charging operating at a fundamental frequency in the range from80 kHz to 90 kHz, magnetic flux densities in the inductive power region(e.g., above a primary wireless power transfer structure) can reachrelatively high levels (e.g., above 2 mT) to allow for sufficient powertransfer (e.g., 3.3 kW, 7 kW, 11 kW, and the like). Therefore, metallicobjects or other objects present in the magnetic field can experienceundesirable induction heating. For this reason, foreign object detection(FOD) may be implemented to detect metallic objects or other objectsthat are affected by the magnetic field generated by the primary and/orthe secondary wireless power transfer structure of the inductive WPTsystem.

In certain applications for inductive wireless charging of electricvehicles, it may also be useful to be able to detect living objects thatmay be present within or near an inductive power region where the levelof electromagnetic field exposure exceeds certain limits (e.g., asdefined by the International Commission on Non-Ionizing RadiationProtection (ICNIRP) recommendation). For this reason, living objectdetection (LOD) may be implemented to detect living objects (e.g., humanextremities, animals), or other objects that may be exposed to themagnetic field generated by the primary and/or the secondary wirelesspower transfer structure of the inductive WPT system.

In further applications for inductive wireless charging of electricvehicles, it may also be useful to be able to detect a vehicle or thetype of vehicle that may be present above the wireless power transmitter(e.g., above the primary wireless power transfer structure). For thisreason, vehicle detection (VD) may be implemented. In yet anotherapplication for inductive wireless charging of electric vehicles, it mayalso be useful to be able to transmit data (e.g., a vehicle identifieror the like) from the vehicle-based secondary device to the ground-basedprimary device. For this reason, vehicle detection (VD) may be extendedfor receiving low rate signaling from the vehicle.

Efficiency of an inductive WPT system for electric vehicle chargingdepends at least in part on achieving sufficient alignment between theground-based primary wireless power transfer structure and the secondarywireless power transfer structure. Therefore, in certain applicationsfor inductive wireless charging of electric vehicles, it may be usefulto be able to determine a position of the vehicle relative to thewireless power transmitter for purposes of guidance and alignment. Morespecifically, it may be useful to be able to determine a position of thevehicle-based wireless power transfer structure (e.g., the secondarywireless power transfer structure) relative to the ground-based wirelesspower transfer structure (e.g., the primary wireless power transferstructure). For this reason, position determination (PD) may beimplemented.

In an aspect of hardware complexity reduction and cost saving, it may beuseful and desirable to provide FOD, LOD, VD, and PD by a commonmulti-purpose detection circuit.

SUMMARY

In one aspect of the disclosure, an apparatus for determining at leastone of a presence of a metallic object, living object, vehicle, type ofvehicle, and a vehicle position is provided. The apparatus includes aplurality of inductive sense circuits and a plurality of capacitivesense circuits. Each of the plurality of inductive sense circuitsincludes at least one inductive sense element (e.g., a sense coil) andan associated capacitive element to compensate for the gross reactanceas presented at the terminals of the at least one inductive senseelement at an operating frequency herein referred to as the sensefrequency. Each of the plurality of capacitive sense circuits includesat least one capacitive sense element (e.g., a sense electrode) and anassociated inductive element to compensate for the gross reactance aspresented at the terminals of the at least one capacitive sense elementat the sense frequency. At least one of the plurality of inductive andcapacitive sense circuits also includes an impedance matching element(e.g., a transformer) for transforming the impedance of the sensecircuit to match with an operating impedance range of the apparatus. Theapparatus further includes a measurement circuit for selectively andsequentially measuring an electrical characteristic (e.g., an impedance)in each of the plurality of inductive and capacitive sense circuitsaccording to a predetermined time multiplexing scheme. Morespecifically, the measurement circuit includes a driver circuitincluding multiplexing (input multiplexing) electrically connected tothe plurality of inductive and capacitive sense circuits for selectivelyand sequentially driving each of the plurality of sense circuits with adrive signal (e.g., a current signal) at the sense frequency based on adriver input signal. The measurement circuit further includes ameasurement amplifier circuit including multiplexing (outputmultiplexing) electrically connected to the plurality of inductive andcapacitive sense circuits for selectively and sequentially amplifying ameasurement signal (e.g., a voltage signal) in each of the plurality ofsense circuits and for providing a measurement amplifier output signalindicative of the measurement signal in each of the plurality of sensecircuits. The measurement circuit also includes a signal generatorcircuit electrically connected to the input of the driver circuit forgenerating the driver input signal. The measurement circuit furtherincludes a signal processing circuit electrically connected to theoutput of the measurement amplifier circuit for receiving and processingthe measurement amplifier output signal and for determining theelectrical characteristic in each of the plurality of inductive andcapacitive sense circuits based on the driver input signal and themeasurement amplifier output signal. The apparatus further includes acontrol and evaluation circuit electrically connected to the measurementcircuit for controlling the signal generator circuit, for controllingthe input and output multiplexing according to the predetermined timemultiplexing scheme, for evaluating the electrical characteristic asmeasured in each of the inductive and capacitive sense circuits, and fordetermining at least one of a presence of a metallic object, livingobject, vehicle, type of vehicle, and a vehicle position based onchanges in the measured electrical characteristics.

In another aspect of the disclosure, a method for determining at leastone of a presence of a metallic object, living object, vehicle, type ofvehicle, and a vehicle position is provided. The method includesselectively and sequentially measuring, in a measurement circuit, anelectrical characteristic (e.g., an impedance) in each of the pluralityof inductive and capacitive sense circuits according to a predeterminedtime multiplexing scheme. More specifically, the method includesselectively and sequentially applying, from a driver circuit as part ofthe measurement circuit and including input multiplexing, a drive signal(e.g., a current signal) at a sense frequency to each of the pluralityof inductive and capacitive sense circuits according to thepredetermined time multiplexing scheme. The method further includesselectively and sequentially amplifying, in a measurement amplifiercircuit as part of the measurement circuit, and including outputmultiplexing, a measurement signal (e.g., a voltage signal) in each ofthe plurality of inductive and capacitive sense circuits according tothe predetermined time multiplexing scheme, and providing a measurementamplifier output signal indicative for the measurement signal. Themethod further includes applying, from a signal generator circuit aspart of the measurement circuit, a driver input signal to the drivercircuit. The method further includes receiving and processing, in asignal processing circuit as part of the measurement circuit, themeasurement amplifier output signal, and determining the electricalcharacteristic in each of the plurality of inductive and capacitivesense circuits based on the driver input signal and the measurementamplifier output signal. The method further includes controlling, in acontrol and evaluation circuit, the signal generator circuit and theinput and output multiplexing according to the time multiplexing scheme.The method further includes evaluating the electrical characteristic asmeasured in each of the inductive and capacitive sense circuits anddetermining at least one of a presence of a metallic object, livingobject, vehicle, type of vehicle, and a vehicle position based onchanges in the measured electrical characteristics.

BRIEF DESCRIPTION OF THE DRAWINGS

In the figures, the third and fourth digit of a reference numberidentify the figure in which the reference number first appears. The useof the same reference numbers in different instances in the descriptionor the figures indicates like elements.

FIG. 1 is a schematic view illustrating an example implementation of amulti-purpose detection circuit including a plurality of inductive andcapacitive sense circuits, a non-living (e.g., metallic) object, and aliving object.

FIG. 2 is a schematic view illustrating an example implementation of awireless power transfer structure of a wireless power transmitterintegrating a portion of the multi-purpose detection circuit shown inFIG. 2, the non-living and the living object of FIG. 1.

FIG. 3 is a vertical cut view illustrating a portion of a WPT systemincluding the vehicle-based wireless power transfer structure and theground-based wireless power transfer structure integrating a portion ofthe multi-purpose detection circuit of FIG. 1, and the non-living andthe living object of FIG. 1.

FIG. 4 is a generic block diagram of an example implementation of themulti-purpose detection circuit o FIG. 1.

FIG. 5A is a schematic diagram of a circuit illustrating an exampleimplementation of a portion of the multi-purpose detection circuit ofFIG. 1 based on inductive sensing and an impedance measurement approach,and the non-living and the living object of FIG. 1.

FIG. 5B is a schematic diagram of a circuit illustrating another exampleimplementation of a portion of the multi-purpose detection circuit ofFIG. 1 based on inductive sensing and the impedance measurement approachof FIG. 5A, and the non-living and the living object of FIG. 1.

FIG. 5C is a schematic diagram of a circuit illustrating an exampleimplementation of a portion of the multi-purpose detection circuit ofFIG. 1 based on inductive sensing and another impedance measurementapproach, and the non-living and the living object of FIG. 1.

FIG. 5D is a schematic diagram of a circuit illustrating an exampleimplementation of a portion of the multi-purpose detection circuit ofFIG. 1 based on inductive sensing and a transimpedance measurementapproach, and the non-living and the living object of FIG. 1.

FIG. 5E is a schematic diagram of a circuit illustrating another exampleimplementation of a portion of the multi-purpose detection circuit ofFIG. 1 based on inductive sensing and the transimpedance measurementapproach of FIG. 5D, and the non-living and the living object of FIG. 1.

FIG. 5F illustrates an equivalent circuit model of the exampleimplementation of FIG. 5A.

FIG. 5G illustrates an equivalent circuit model of the exampleimplementation of FIG. 5C.

FIG. 5H illustrates an equivalent circuit model of a portion of thecircuits of FIGS. 5C, 7C, 7F, and 7H.

FIG. 5I illustrates an equivalent circuit model of another portion ofthe circuits of FIGS. 5D and 5E.

FIG. 5J illustrates another equivalent circuit model of the portion ofthe circuits of FIGS. 5D and 5E illustrated in FIG. 5I.

FIG. 5K shows a table of equations that may be relevant for theequivalent circuit models of FIG. 5F and FIG. 5G.

FIG. 6 illustrates a complex impedance plane, different types of objectsof FIG. 1, and corresponding areas where changes of impedance may occurin presence of the object.

FIG. 7A is a schematic diagram of a circuit illustrating an exampleimplementation of a portion of the multi-purpose detection circuit ofFIG. 1 based on capacitive sensing and the impedance measurementapproach of FIG. 5A, and the living and the non-living object of FIG. 1.

FIG. 7B is a schematic diagram of a circuit illustrating another exampleimplementation of a portion of the multi-purpose detection circuit ofFIG. 1 based on capacitive sensing and the impedance measurementapproach of FIG. 5A, and the living and the non-living object of FIG. 1.

FIG. 7C is a schematic diagram of a circuit illustrating a furtherexample implementation of a portion of the multi-purpose detectioncircuit of FIG. 1 based on capacitive sensing and the impedancemeasurement approach of FIG. 5A, and the living and the non-livingobject of FIG. 1.

FIG. 7D is a schematic diagram of a circuit illustrating yet anotherexample implementation of a portion of the multi-purpose detectioncircuit of FIG. 1 based on capacitive sensing and the impedancemeasurement approach of FIG. 5A, and the living and the non-livingobject of FIG. 1.

FIG. 7E is a schematic diagram of a circuit illustrating an exampleimplementation of a portion of the multi-purpose detection circuit ofFIG. 1 based on capacitive sensing and the impedance measurementapproach of FIG. 5C, and the living and the non-living object of FIG. 1.

FIG. 7F is a schematic diagram of a circuit illustrating another exampleimplementation of a portion of the multi-purpose detection circuit ofFIG. 1 based on capacitive sensing and the impedance measurementapproach of FIG. 5C, and the living and the non-living object of FIG. 1.

FIG. 7G is a schematic diagram of a circuit illustrating an exampleimplementation of a portion of the multi-purpose detection circuit ofFIG. 1 based on capacitive sensing and the transimpedance measurementapproach of FIG. 5D, and the living and the non-living object of FIG. 1.

FIG. 7H is a schematic diagram of a circuit illustrating another exampleimplementation of a portion of the multi-purpose detection circuit ofFIG. 1 based on capacitive sensing and the transimpedance measurementapproach of FIG. 5D, and the living and the non-living object of FIG. 1.

FIG. 7I is a schematic diagram of a circuit illustrating a furtherexample implementation of a portion of the multi-purpose detectioncircuit of FIG. 1 based on capacitive sensing and the transimpedancemeasurement approach of FIG. 5D, and the living and the non-livingobject of FIG. 1.

FIG. 7J illustrates an equivalent circuit model of the exampleimplementation of FIG. 7A.

FIG. 7K illustrates an equivalent circuit model of the exampleimplementation of FIG. 7E.

FIG. 7L illustrates an equivalent circuit model of a portion of thecircuits of FIGS. 7G, 7H, and 7I.

FIG. 7M illustrates another equivalent circuit model of a portion of thecircuits of FIGS. 7G, 7H, and 7I.

FIG. 7N shows a table of equations that may be relevant for theequivalent circuit models of FIG. 7J and FIG. 7K.

FIG. 8A illustrates a complex impedance plane, different types ofobjects of FIG. 1, and corresponding areas where changes of impedancemay occur in presence of the object.

FIG. 8B illustrates an equivalent circuit model applicable to an objectof FIG. 1 proximate to the capacitive sense element of FIG. 7A.

FIG. 8C shows a normalized admittance chart indicating lines of constantreal permittivity and constant imaginary permittivity.

FIG. 8D shows the normalized admittance chart of FIG. 8C indicatingmeasured admittance changes in presence of an object of FIG. 1.

FIG. 8E shows another normalized admittance chart indicating the angleof a portion of the measured admittance changes of FIG. 8D.

FIG. 8F shows the normalized admittance chart of FIG. 8E indicating theangle of another portion of the measured admittance changes of FIG. 8D.

FIG. 8G shows a diagram indicating a normalized effective conductivityand susceptibility as determined from the measured admittance changes ofFIG. 8D.

FIG. 8H illustrates a complex plane indicating an effective complexpermittivity as determined from the measured admittance changes of FIG.8D.

FIG. 9A is a schematic diagram of a circuit illustrating an exampleimplementation of a portion of the multi-purpose detection circuit ofFIG. 1 including a plurality of inductive and capacitive sense circuits.

FIG. 9B is a schematic diagram illustrating an example implementation ofa portion of the circuit of FIG. 9A.

FIG. 9C is a schematic diagram illustrating an example implementation ofanother portion of the circuit of FIG. 9A.

FIG. 9D is a schematic diagram illustrating an example implementation ofa further portion of the circuit of FIG. 9A.

FIG. 10 is a schematic diagram of a circuit illustrating another exampleimplementation of a portion of the multi-purpose detection circuit ofFIG. 1 including a plurality of inductive and capacitive sense circuits.

FIG. 11 is a schematic diagram of a circuit illustrating a furtherexample implementation of a portion of the multi-purpose detectioncircuit of FIG. 1 including a plurality of inductive and capacitivesense circuits.

FIG. 12A is a schematic view illustrating an example implementation ofthe housing of the ground-based wireless power transfer structureintegrating single-ended capacitive sense elements of the multi-purposedetection circuit of FIG. 1.

FIG. 12B is a schematic view illustrating an example implementation ofthe housing of the ground-based wireless power transfer structureintegrating double-ended capacitive sense element of the multi-purposedetection circuit of FIG. 1.

FIG. 13A is a schematic view illustrating an example printed circuitboard implementation of a holohedral sense electrode.

FIG. 13B is schematic view illustrating an example printed circuit boardimplementation of a sense electrode having a finger structure.

FIGS. 14A to 14C illustrates an electric vehicle approaching aground-based wireless power transfer structure installed in a parkingspace.

FIGS. 15A and 15B illustrates an example implementation of vehicleposition determination (PD) based on pattern detection.

DETAILED DESCRIPTION

The detailed description set forth below in connection with the appendeddrawings is intended as a description of example implementations and isnot intended to represent the only implementations in which thetechniques described herein may be practiced. The term “example” usedthroughout this description means “serving as an example, instance, orillustration,” and should not necessarily be construed as preferred oradvantageous over other example implementations. The detaileddescription includes specific details for the purpose of providing athorough understanding of the example implementations. In someinstances, some devices are shown in block diagram form. Drawingelements that are common among the following figures may be identifiedusing the same reference numerals.

As mentioned above foreign object detection (FOD) (and particularlymetal object detection) may be valuable for a variety of applications.For detection in a predetermined region, a FOD system may include aplurality of inductive sense circuits each including an inductive senseelement (e.g., a sense coil) distributed across a predetermined area(e.g., a planar array of sense coils integrated into the ground-basedwireless power transfer structure). The predetermined region may bedefined by the space where metal objects may be found and where themagnetic flux density exceeds certain limits (e.g., a thresholddetermined based on what levels of temperature a metal object might beheated up). This is generally a three-dimensional space above theplurality of indictive sense elements. The number of the inductive senseelements may be proportional or related to the minimum size of objectsthat are desirable to be detected. For a system that is configured todetect small objects (e.g., a paper clip), the number of sense elementsmay be relatively high (e.g., in the order of 100). An example FODsystem is described in U.S. Pat. No. 10,627,257, titled Systems,Methods, and Apparatus for Detection of Metal Objects in a PredeterminedSpace, the entire contents of which are hereby incorporated byreference.

As mentioned above living object detection (LOD) (e.g., humanextremities, animals) may be valuable for a variety of applications. Fordetection in a predetermined region, a LOD system may include aplurality of capacitive sense circuits each including a capacitive senseelement (e.g., a sense electrode) e.g., disposed along the periphery ofa ground-based wireless power transfer structure of a WPT system. Thepredetermined region may be defined by the space accessible for livingobjects and where living objects may be located and where the exposuremagnetic field strength exceeds certain limits (e.g., as recommended byICNIRP). This is generally a three-dimensional space. The number of thecapacitive sense elements may be proportional or related to the minimumsize of living objects that are desirable to be detected. For a systemthat is configured to detect human extremities (e., a hand) and animals(e.g., a cat), the number of sense elements may be relatively low (e.g.,in the order of 4). A measurement drive circuitry for applying drivesignals to each of the plurality of capacitive sense circuits eachincluding a capacitive sense element and additional elements forconditioning, as well as corresponding measurement circuitry as neededfor measuring an electrical characteristic in each of the plurality ofcapacitive sense circuits and for looking for changes in the electricalcharacteristics that may correspond to the presence of a living object.An example LOD system is described in U.S. Pat. No. 9,952,266, titledObject Detection for Wireless Energy Transfer Systems, the entirecontents of which are hereby incorporated by reference.

As mentioned above vehicle detection (VD) or detection of the type ofvehicle above the ground-based wireless power transfer structure of aWPT system may be valuable for a variety of applications. For detectionof a vehicle or the type of vehicle, a VD system may include a pluralityof inductive sense circuits each including an inductive sense element(e.g., a sense coil) distributed across an area defined by theground-based wireless power transfer structure (e.g., a planar array ofsense coils) and a plurality of capacitive sense circuits each includinga capacitive sense element (e.g., a sense electrode) disposed in an areadefined by the ground-based wireless power transfer structure. Drivecircuitry for applying drive signals to each of the inductive andcapacitive sense circuits, each including an inductive and capacitivesense element, respectively and additional elements for conditioning, aswell as corresponding measurement circuitry as needed for measuring anelectrical characteristic in each of the plurality of capacitive sensecircuits and for looking for changes in the electrical characteristicsthat may correspond to the presence of a vehicle.

As mentioned above determination of a position (PD) of a vehicle (e.g.,the position of the vehicle-based wireless power transfer structurerelative to the ground-based wireless power transfer structure of a WPTsystem) may be valuable for a variety of applications. For determinationof a vehicle position, a PD system may include a plurality of inductivesense circuits each including an inductive sense element (e.g., a sensecoil) distributed across an area defined by the ground-based wirelesspower transfer structure (e.g., a planar array of sense coils) and aplurality of capacitive sense circuits each including a capacitive senseelement (e.g., a sense electrode) disposed in an area defined by theground-based wireless power transfer structure.

In some implementations, the PD system is configured to support apassive beacon PD technique. Passive beacon PD uses at least one passivebeacon transponder that may be integrated into the vehicle-basedwireless power transfer structure or that may be mounted elsewhere atthe vehicle underbody. When positioned above the inductive andcapacitive sense element array of the multi-purpose detection circuit,the passive beacon transponder produces a distinct time-varying change(a modulated response) in the electrical characteristic of at least oneof the plurality of inductive sense circuits and capacitive sensecircuits. This modulated response may be used for determining a positionof the at least one passive beacon transponder relative to the array ofsense elements, which is related to the position of the vehicle-basedwireless power transfer structure relative to the ground-based wirelesspower transfer structure. The at least one passive beacon transpondermay also be used for determining presence of a vehicle (VD) or the typeof vehicle e.g., by means of a modulation that is characteristic for thetype of vehicle. Further, the at least one passive beacon transpondermay be used to transmit data (e.g., at a low data rate) to the primarydevice by means of the passive modulation technique.

In some implementations, the at least one passive beacon transponderincludes an inductive passive beacon transponder configured to mainlyinteract with the inductive sense circuits. In other implementations,the at least one passive beacon transponder includes a capacitivepassive beacon transponder configured to mainly interact with thecapacitive sense circuits. In further implementations, the at least onepassive beacon transponder is configured to interact with both theinductive and capacitive sense circuits. An example inductive passivebeacon PD system is described in U.S. patent application Ser. No.16/052,445, titled Hybrid Foreign Object Detection and PositioningSystem, the entire contents of which are hereby incorporated byreference.

Circuitry for applying drive signals to each of the plurality ofinductive and/or capacitive sense circuits each including a senseelement and additional elements for conditioning, as well ascorresponding measurement, control and evaluation circuitry as neededfor measuring an electrical characteristic in each of the plurality ofinductive sense circuits and detecting changes in the electricalcharacteristics that may be indicative of one of the presence of a metalobject, a living object, a vehicle, the type of vehicle, and a vehicleposition may be complex and costly as the number of sense elementsincreases. Therefore, in an aspect of hardware complexity reduction andcost saving, it may be useful and desirable to combine the variousfunctions such as FOD, LOD, VD, data signaling, and PD in a singlesystem referred to herein as the multi-purpose detection circuit.

An electric vehicle is used herein to describe a remote system, anexample of which is a vehicle that includes, as part of its locomotioncapabilities, electrical power derived from a chargeable energy storagedevice (e.g., one or more rechargeable electrochemical cells or othertype of battery). As non-limiting examples, some electric vehicles maybe hybrid electric vehicles that include, besides electric motors, atraditional combustion engine for direct locomotion or to charge thevehicle's battery. Other electric vehicles may draw all locomotionability from electrical power. An electric vehicle is not limited to anautomobile and may include motorcycles, carts, scooters, and the like.

A foreign object is used herein to describe an object that does notnaturally belong to the WPT system. A foreign object may include ametallic object, a non-living dielectric (substantially nonconductive)object, a living object (e.g., an animal, a human extremity), a vehicle,or a combination thereof. It may describe an object that needs to bedetected for purposes of safety of equipment or persons, but it may alsorefer to an object of no harm that is potential to produce a falsepositive detection in a multi-purpose detection system.

FIG. 1 illustrates an example implementation of a multi-purposedetection circuit 100 that includes a plurality of inductive sensecircuits 106 and a plurality of capacitive sense circuits 108illustrated in FIG. 1 by inductive sense circuits 106 a, 106 b, somedots, and 106 n and by capacitive sense circuits 108 a, 108 b, somedots, and 108 n. The dots shall indicate that the number of inductivesense circuits 106 and/or the number of capacitive sense circuits 108may be greater than three. The plurality of inductive sense circuits 106is also sometimes referred herein as the plurality of inductive sensecircuits 106 a, 106 b, . . . , 106 n. Likewise, the plurality ofcapacitive sense circuits 108 is also sometimes referred herein as theplurality of capacitive sense circuits 108 a, 108 b, . . . , 108 n. Asillustrated in FIG. 1, each of the inductive sense circuit of theplurality of sense circuits 106 a, 106 b, . . . , 106 n includes acorresponding inductive sense element (e.g., a sense coil) of aplurality of inductive sense elements 107 a, 107 b, . . . , 107 n,respectively. Likewise, each of the capacitive sense circuits of theplurality of sense circuits 108 a, 108 b, 108 n includes a correspondingcapacitive sense element (e.g., a pair of sense electrodes) of aplurality of capacitive sense elements 109 a, 109 b, . . . , 109 n,respectively.

FIG. 1 also illustrates foreign objects 110 and 112 as referred toherein as non-living objects and a living object 114. The object 110 mayrepresent a metallic (substantially electrically conductive object) thatis potentially heated when exposed to the WPT magnetic field aspreviously discussed, while the object 112 may be representative for adielectric or ferromagnetic object that is substantially electricallynon-conductive and that does not heat to hazardous temperatures whenexposed to the WPT magnetic field. The living object 114 may stand for ahuman extremity (e.g., a hand as depicted in FIG. 1) or an animal thatis dielectric and substantially electrically non-conductive.

The inductive sense elements 107 a, 107 b, . . . , 107 n and capacitivesense elements 109 a, 109 b, . . . , 109 n are configured to sense atleast one of a presence of a foreign object (e.g., object 110) inproximity to at least one of the plurality of inductive sense elements107 a, 107 b, . . . , 107 n, a living object (e.g., object 114) inproximity to at least one of the plurality of capacitive sense elements109 a, 109 b, 10 n, a vehicle or type of vehicle (not shown in FIG. 1)positioned above the plurality of inductive and capacitive senseelements 107 a-107 n and 109 a-109 n, respectively, and for determininga vehicle position based on measuring one or more electricalcharacteristics (e.g., an impedance) in each of the plurality ofinductive sense circuits 106 a, 106 b, . . . , 106 n and capacitivesense circuits 108 a, 108 b, . . . , 108 n and based on detectingchanges in the measured one or more electrical characteristics. Each ofthe plurality of inductive sense circuits 106 a, 106 b, . . . , 106 nand capacitive sense circuits 108 a, 108 b, . . . , 108 n may alsoinclude additional conditioning circuitry (not shown in FIG. 1) e.g.,configured to improve measurement of the one or more electricalcharacteristics and thus sensitivity and reliability of themulti-purpose detection circuit 100. Each of the plurality of sensecircuits also defines at least one measurement port (not shown inFIG. 1) where the one or more electrical characteristics is measured andrefers to.

Each of the plurality of inductive sense elements 107 a, 107 b, . . . ,107 n is shown in FIG. 1 as a “circular” coil for purposes ofillustration. However, in other implementations, the inductive senseelements 107 a, 107 b, . . . , 107 n may include a sense coil havinganother coil topology, e.g., a “figure-eight-like” topology. In yetother implementations, the plurality of inductive sense elements 107 a,107 b, . . . , 107 n, may include sense coils of a mixed coil topology,e.g., “circular” and “figure-eight-like”. In further implementations,the plurality of inductive sense elements 107 a, 107 b, . . . , 107 n,may include sense coils (e.g., solenoid coils) with a ferrite core (notshown herein) that are physically smaller compared to “air” coils. Inyet further implementations, the plurality of sense elements 107 a, 107b, . . . , 107 n may include other inductive devices that can be usedfor generating a magnetic field for detecting a foreign object (e.g.,object 110), a vehicle, or for determining a vehicle position. In someimplementations (not shown herein), each of the plurality of inductivesense elements 107 a, 107 b, . . . , 107 n, may include a double or evena triple sense coil arrangement that may be used in conjunction with atransimpedance or mutual impedance measurement technique. In someimplementations, the plurality of inductive sense elements 107 a, 107 b,. . . , 107 n is arranged in an array 107, such as a two-dimensionalarray 107 as shown in FIG. 1. However, in other implementations, thesense elements of the plurality of inductive sense elements 107 a, 107b, . . . , 107 n are arranged in other configurations that do notconform to rows or columns (radial or interleaved), are at leastpartially overlapping or have irregular spacing, have different size,have different shapes (circular, hexagonal, etc.), or cover irregulardetection areas, or any combination thereof. As such the term “array” asused herein denotes a plurality of sense elements that are arranged overa predetermined area. Furthermore, the number of sense elements of anarray 107 and thus the number of sense circuits can vary widely based onthe application including the total region in which a foreign object(e.g., object 110) is to be detected and the smallest size of an objectthe multi-purpose detection circuit 100 is configured to detect. Exampleimplementations of the inductive sense element (e.g., 107 a) andarrangements of inductive sense elements are described in U.S. Pat. No.9,726,518, titled Systems, Methods, and Apparatus for Detection of MetalObjects in a Predetermined Space, in U.S. patent application Ser. No.16/358,534, titled Foreign Object Detection Circuit Using MutualImpedance Sensing, in U.S. Pat. No. 10,122,192, titled Sense CoilGeometries with Improved Sensitivity for Metallic Object Detection in aPredetermined Space, in U.S. Pat. No. 10,124,687, titled Hybrid ForeignObject Detection (FOD) Loop Array Board, the entire contents of whichare hereby incorporated by reference.

Each of the plurality of capacitive sense elements 109 a, 109 b, . . . ,109 n is shown in FIG. 1 as a pair of sense electrodes for purposes ofillustration. However, in other implementations, the capacitive senseelements 109 a, 109 b, 10 n may include a single electrode providing asingle terminal. In further implementations, the capacitive senseelements 109 a, 109 b, . . . , 109 n, may be driven and configured formeasuring a transimpedance (a mutual capacitance). In yet furtherimplementations, the capacitive sense elements 109 a, 109 b, 109 n, mayinclude other capacitive devices that can be used for generating anddetecting an electric field for detecting a foreign object (e.g., object112), a living object (e.g., object 114), a vehicle (e.g., vehicle 330),or for determining a type of vehicle or a vehicle position. In FIG. 1,the capacitive sense elements 109 a, 109 b, . . . , 109 n, are shownarranged in an area around the array of inductive sense elements 107 a,107 b, . . . , 107 n. However, in other implementations, the capacitivesense elements of the plurality of capacitive sense elements 109 a, 109b, . . . , 109 n are arranged in other configurations, e.g., distributedover the area of the array 107 of the inductive sense elements. Exampleimplementations of the capacitive sense element (e.g., 109 a) andarrangements of capacitive sense elements are described in U.S. Pat. No.9,952,266, titled Object Detection for Wireless Energy Transfer Systems,the entire contents of which are hereby incorporated by reference.

Each of the plurality of inductive sense circuits 106 and the pluralityof capacitive sense circuits 108 including a corresponding sense elementof the plurality of inductive sense elements 107 a-107 n and theplurality of capacitive sense elements 109 a-109 n are operablyconnected to a measurement circuit 104. The measurement circuit 104,including multiplexing (not shown in FIG. 1), is configured toselectively and sequentially measure one or more electricalcharacteristics in each of the plurality of inductive and capacitivesense circuits (106 and 108, respectively) and to provide outputs to thecontrol and evaluation circuit 102.

The measurement circuit 104 is configured to cause each of the pluralityof inductive sense elements (e.g., sense coils) 107 a, 107 b, . . . ,107 n to selectively and sequentially generate an alternating magneticfield at the sense frequency, e.g., by selectively and sequentiallyapplying a sense signal (e.g., a current) to each of the plurality ofinductive sense circuits 106 a, 106 b, . . . , 106 n. If a metallicobject (e.g., object 110) is present in the alternating magnetic field,eddy currents will be generated in the object. According to Lentz' law,the eddy currents in the object will generate another (secondary)magnetic field that interacts with the primary magnetic field asgenerated by the respective sense element, and a mutual coupling isdeveloped. This may cause a change in an electrical characteristic(e.g., an impedance) as measured by the measurement circuit 104 in therespective inductive sense circuit (e.g., inductive sense circuit 106a). A change in a measured electrical characteristic may also be causedby a substantially non-conductive but ferromagnetic object (e.g., object112) with a relative permeability μ_(r)>1 that interacts with thealternating magnetic field as generated by the respective sense element.Applying a sense signal to an inductive sense circuit (e.g., sensecircuit 106 a) may also cause the respective inductive sense element togenerate an alternating electric field that may interact with asubstantially non-conductive, dielectric object (e.g., non-living object112 or living object 114) causing a change in the electricalcharacteristic as measured in the respective inductive sense circuit(capacitive sensing effect). This alternating electric field may alsointeract with a metallic (substantially electrically conductive) object(e.g., object 110). However, this effect may be orders of magnitudeweaker than the magnetic field effect.

The measurement circuit 104 is further configured to cause each of theplurality of capacitive sense elements (e.g., sense electrodes) 109 a,109 b, . . . , 109 n to selectively and sequentially generate analternating electric field at the sense frequency, e.g., by selectivelyand sequentially applying a sense signal (e.g., a current) to each ofthe plurality of capacitive sense circuits 108 a, 108 b, . . . , 108 n.If a substantially non-conductive, dielectric object (e.g., livingobject 114 or non-living object 112) with a relative permittivityε_(r)>1 is present in the alternating electric field, it will interactwith the electric field. This may cause a change in an electricalcharacteristic (e.g., an impedance) as measured by the measurementcircuit 104 in the respective capacitive sense circuit (e.g., capacitivesense circuit 108 a). A change in a measured electrical characteristicmay also be caused by a metallic object (e.g., object 110) as it willalso interact with the alternating electric field as generated by therespective capacitive sense element. Applying a sense signal (e.g.,current) to a capacitive sense circuit (e.g., sense circuit 106 a) mayalso cause the respective capacitive sense element to generate analternating magnetic field that may interact with a metallic object(e.g., object 110) causing a change in the electrical characteristic asmeasured in the respective capacitive sense circuit (inductive sensingeffect). However, this effect may be orders of magnitude weaker than theelectric field effect.

The control and evaluation circuit 102 is configured to control themeasurement circuit 104 (e.g., the multiplexing) and to evaluate theoutputs of the measurement circuit 104, to determine at least one of apresence of a foreign object (e.g., object 110), living object (e.g.,object 114), a presence of a vehicle with reference to FIG. 3, a type ofvehicle, and a vehicle position based on changes in the measured one ormore electrical characteristics. In some implementations, the controland evaluation circuit 102 may include the decision functions as neededfor FOD, LOD, and VD as well as the position calculation functionsneeded for PD. In other implementations, the vehicle position isdetermined in a unit external to the multi-purpose detection circuit 100(not shown herein) based on outputs (e.g., raw data) from the controland evaluation circuit 102 and on outputs provided by other ground- orvehicle-based sensors (not shown herein).

FIG. 2 illustrates an example implementation of a wireless powertransfer structure 200 that is a portion of a WPT system including aportion of the multi-purpose detection circuit 100 of FIG. 1. Thewireless power transfer structure 200 may depict either a wireless powertransmitter that generates a magnetic field (e.g., at an operatingfrequency in the range from 80-90 kHz) for transferring power or awireless power receiver that can couple and receive power via a magneticfield. It may be more likely that when integrated with a multi-purposedetection circuit 100, the wireless power transfer structure 200 may bea wireless power transmitter as power may be generally transferred fromthe ground or other upward facing surface where foreign objects (e.g.,object 110 or 112) will generally come to a rest. However otherimplementations are possible, e.g., the multi-purpose detection circuit100 or a portion thereof may be also integrated into a wireless powerreceiver (e.g., a vehicle-based wireless power transfer structure). Thewireless power transfer structure 200 (also sometimes referred to as a“ground assembly” or “base bad”) may be configured to wirelesslytransmit or receive power.

The wireless power transfer structure 200 includes a coil 202 (e.g., aLitz wire coil) also referred to as the WPT coil that is configured togenerate an alternating magnetic field when driven with a current by apower conversion circuit (not shown herein). The wireless power transferstructure 200 may further include a ferrite 204 structure configured tochannel and/or provide a path for magnetic flux (e.g., may be arrangedin one or more ferrite tiles). The wireless power transfer structure 200may also include a metal shield 206 (also sometimes referred to as aback plate). The metal shield 206 is configured to prevent the magneticfield or associated electromagnetic emissions from extending far beyonda boundary determined by the shield 206 or at least to attenuate themagnetic field extending beyond that boundary. As an example, the shield206 may be formed from aluminum.

FIG. 2 illustrates one example how the plurality of inductive senseelements (array 107) and the plurality of capacitive sense elements 109of FIG. 1 may be integrated into the wireless power transfer structure200.

FIG. 3 illustrates a vertical cut view of a portion 300 of a WPT systemapplicable to wireless electric vehicle charging. This portion 300includes the ground-based (e.g., transmit) wireless power transferstructure 200 with reference to FIG. 2 and the vehicle-based (e.g.,receive) wireless power transfer structure 310. The ground-basedwireless power transfer structure 200 includes the shield (back plate)206, a layer of ferrite 204, and a WPT coil 202 with reference to FIG.2. It also includes a housing 328 configured to house the WPT coil 202,the ferrite 204, and the shield 206. In addition, the housing 328 isconfigured to house the plurality of inductive sense elements (array107) and the plurality of capacitive sense elements (109) as part of themulti-purpose detection circuit 100 as illustrated in FIG. 2. In someimplementations, the shield 206 may form a portion of the housing 328 asillustrated in FIG. 3. Further, the housing 328 may be inclined alongits perimeter from its edge toward its interior to form a ramp overwhich a vehicle may drive. The power conversion circuit (not shownherein) may be electrically connected to the WPT coil 202 or a portionor all may also be housed in the housing 328. In some aspects, thecapacitive sense elements (e.g., the capacitive sense elements 109 a,109 b, . . . , 109 n) may be oriented to be nonparallel with a planedefined by the array 107 of inductive sense elements. For example, thecapacitive sense elements may be oriented to be substantially parallelto the inclined top surface of the housing 328 along the housing'sperimeter.

The vehicle-based wireless power transfer structure 310 includes a WPTcoil 312, a layer of ferrite 315, and a shield 316 made of anelectrically conductive material. In some implementations, the shield316 may be formed from a portion of the apparatus that the ferrite 315and the WPT coil 312 are affixed to the metallic underbody of a vehicle330. In this case, a housing 318 configured to house the WPT coil 312and ferrite 315 is provided but that may not house the shield 316.However other implementations are possible where a conductive back plateis included in the housing 318. A power conversion circuit (not shownherein) may be electrically connected to the WPT coil 312 or a portionor all may also be housed in the housing 318.

As mentioned above and as illustrated in FIG. 3, the vehicle-basedwireless power transfer structure 310 may also integrate at least one ofan inductive passive beacon transponder 313 and a capacitive beacontransponder 314 e.g., for purposes of PD and VD as previously discussed.The inductive passive beacon transponder 313 may be configured toprimarily interact with the inductive sense elements e.g., the inductivesense elements 107 a, 107 b, . . . , 107 n. In some implementations, theinductive passive beacon transponder 313 includes a transponder coil, acapacitive element to compensate for the gross reactance of the coil atthe operating (sense) frequency of the multi-purpose detection circuit100, and a passive impedance modulation circuit (these elements notshown in herein). The capacitive passive beacon transponder 314 may beconfigured to primarily interact with the capacitive sense elementse.g., the capacitive sense elements 109 a, 109 b, . . . , 109 n. In someimplementations, the capacitive passive beacon transponder 314 includesa transponder electrode, an inductive element to compensate for thegross reactance of the electrode at the operating (sense) frequency ofthe multi-purpose detection circuit 100, and a passive impedancemodulation circuit (these elements not shown in herein). In furtherimplementations (not shown herein), the passive beacon transponder(e.g., passive beacon transponder 313) is configured to interact withboth the inductive and capacitive sense elements of the multi-purposedetection circuit 100.

The ground-based (e.g., transmit) wireless power transfer structure 200may be configured to generate a magnetic field 232. The vehicle-basedwireless power transfer structure 310 may be configured to inductivelyreceive power via the magnetic field. Furthermore, as the ground-basedwireless power transfer structure 200 may be positioned on a ground orother top facing surface, an object (e.g., object 110 or 112) may cometo rest at the top surface of the housing 328 as illustrated in FIG. 3.The object may thereby be potentially exposed to high levels of magneticflux density if power is being transferred.

FIG. 4 is a generic block diagram illustrating an example implementationor operation of a multi-purpose detection circuit 100. The circuit 100includes the plurality of inductive sense circuits 106 a, 106 b, . . . ,106 n, including the inductive sense elements 107 a, 107 b, . . . , 107n, respectively, the plurality of capacitive sense circuits 108 a, 108b, . . . , 108 n, including the capacitive sense elements 109 a, 109 b,. . . , 109 n, respectively, the measurement circuit 104, and thecontrol and evaluation circuit 102 with reference to FIG. 1.

Each of the plurality of inductive sense circuits 106 may also includean associated capacitive element (not shown herein) to compensate forthe gross reactance as presented at the terminals of the at least oneinductive sense element at the sense frequency. Each of the plurality ofcapacitive sense circuits 108 may also include an associated inductiveelement (not shown herein) to compensate for the gross reactance aspresented at the terminals of the at least one capacitive sense elementat the sense frequency. At least one of the plurality of inductive andcapacitive sense circuits also includes an impedance matching element(e.g., a transformer) for transforming the impedance of the sensecircuit (e.g., sense circuit 108 a) to match with an operating impedancerange of the multi-purpose object detection circuit 100. In an exampleimplementation, each of the plurality of inductive sense circuits 106 isnaturally matched with an operating impedance range without using anadditional impedance matching element. However, the plurality ofcapacitive sense circuits 108 is not naturally matched, and therefore anadditional impedance matching element (e.g., a transformer) is used. Inanother example implementation, it is vice-versa. In a further exampleimplementation, both the plurality of inductive and capacitive sensecircuits 106 and 108, respectively, include an additional impedancematching element.

The measurement circuit 104 is electrically connected to the pluralityof inductive and capacitive sense circuits and configured forselectively and sequentially measuring one or more electricalcharacteristics (e.g., an impedance) in each of the plurality ofinductive and capacitive sense circuits according to a predeterminedtime multiplexing scheme.

The control and evaluation circuit 102 is electrically connected to themeasurement circuit 104 and configured to control time multiplexing(input multiplexer (MUX) control and output MUX control in FIG. 4)according to the predetermined time multiplexing scheme, to evaluate theone or more electrical characteristics as measured in each of theinductive and capacitive sense circuits, and to determine at least oneof a presence of a foreign object (e.g., object 110 or 112), a livingobject (e.g., object 114), a vehicle (e.g., vehicle 330), a type ofvehicle, and a vehicle position based on changes in the measured one ormore electrical characteristics.

The measurement circuit 104 further includes a driver circuit 402, ameasurement amplifier circuit 404, a signal generator circuit 406, and asignal processing circuit 408.

The driver circuit 402 including multiplexing (input multiplexing) iselectrically connected to the plurality of inductive sense circuits 106and the plurality of capacitive sense circuits 108 and configured toselectively and sequentially apply a drive signal (e.g., a currentsignal) at the sense frequency to each of the plurality of inductivesense circuits 106 and the plurality of capacitive sense circuits 108based on a driver input signal generated by the signal generator circuit406.

The measurement amplifier circuit 404 including multiplexing (outputmultiplexing) is electrically connected to the plurality of inductivesense circuits 106 and the plurality of capacitive sense circuits 108and configured to selectively and sequentially amplify a measurementsignal (e.g., a voltage signal) in each the plurality of inductive sensecircuits 106 and the plurality of capacitive sense circuits 108 and toprovide a measurement amplifier output signal indicative of themeasurement signal in each of the plurality of sense circuits.

The signal generator circuit 406 electrically connected to the input ofthe driver circuit 402 is configured to generate the driver inputsignal.

The signal processing circuit 408 electrically connected to the outputof the measurement amplifier circuit 404 is configured to receive andprocess the measurement amplifier output signal and to determine the oneor more electrical characteristics in each of the plurality of inductiveand capacitive sense circuits based on the driver input signal and themeasurement amplifier output signal.

The dashed lines used in FIG. 4 emphasize that the components and theirconfiguration in the driver circuit 402 and the measurement amplifiercircuit 404 are illustrative, and other implementations may have theseor other components configured to selectively and sequentially drive theplurality of sense circuits 106 and 108 with a drive signal and toselectively and sequentially amplify a measurement signal in each of theplurality of sense circuits 106 and 108. Furthermore, while certaincircuit elements are described as connected between other elements, itshould be appreciated that there may be other circuit elements invarious implementations that may also be in between the two elementsdescribed as electrically connected (e.g., other elements interposed).To mention an example of an alternative implementation (not shownherein), multiplexing is common to both the driver circuit 402 and themeasurement amplifier circuit 404.

Example implementations of the measurement circuit 104 and the controland evaluation circuit 102 are described in U.S. Pat. No. 9,726,518,titled Systems, Methods, and Apparatus for Detection of Metal Objects ina Predetermined Space, U.S. Pat. No. 9,921,045, titled Systems, Methods,and Apparatus for Increased Foreign Object Detection Loop ArraySensitivity, in U.S. Pat. No. 10,295,693, titled Systems, Methods, andApparatus for Foreign Object Detection Loop Based on Inductive ThermalSensing, in U.S. Pat. No. 10,302,795, titled Systems, Methods, andApparatus for Detecting Ferromagnetic Objects in a Predetermined Space,in U.S. Pat. No. 10,298,049, titled Systems, Methods, and Apparatus forDetecting Metallic Objects in a Predetermined Space via inductivekinematic Sensing, in U.S. patent application Ser. No. 16/226,156,titled Foreign Object Detection Circuit Using Current Measurement, inU.S. patent application Ser. No. 16/392,464, titled Extended ForeignObject Detection Signal Processing, and in U.S. patent application Ser.No. 16/358,534, titled Foreign Object Detection Circuit Using MutualImpedance Sensing, the entire contents of which are hereby incorporatedby reference.

In an example operation of the multi-purpose detection circuit 100, thesense signal is selectively and sequentially applied to each of theplurality of inductive sense circuits 106 and to each of the pluralityof the capacitive sense circuits 108 according to a time divisionmultiplexing scheme and in a round robin fashion. The sense signal fordriving an inductive sense circuit (e.g., inductive sense circuit 106 a)is applied in a time interval (time slot) allocated to that sensecircuit and has a maximum duration equal or shorter than the duration ofthe time slot. The time frame corresponding to the sum of time slotsallocated to the plurality of inductive sense circuits 106 andcapacitive sense circuits 108 is also referred herein as to the scancycle or to the repetition period.

In an aspect to reduce the duration of the scan cycle, a first sensesignal is selectively and sequentially applied to each of a portion ofthe plurality of inductive sense circuits 106 and capacitive sensecircuits 108 and a second sense signal is concurrently, selectively andsequentially applied to each of the remaining portions of inductive andcapacitive sense circuits. Concurrently applying two or more sensesignals reduces the scan cycle and may result in a reduced detectionlatency with respect to FOD and LOD and in an increased position updaterate with respect to PD (e.g., using the passive beaconing approach aspreviously described).

In an example implementation and operation of the multi-purposedetection circuit 100, the first and the at least one concurrentlyapplied second sense signal are sinusoidal signals of the samefrequency.

In another example implementation and operation of the multi-purposedetection circuit 100, the first and the at least one concurrentlyapplied second sense signal are sinusoidal signals but differ infrequency.

In a further example implementation and operation of the multi-purposedetection circuit 100, each of the first and the at least one concurrentsecond sinusoidal sense signals as applied in time slots allocated tothe same sense circuit (e.g., sense circuit 106 a) start with the samephase (e.g., zero-phase). In some implementations using more than twocurrent sense signals, starting sense signals in time slots allocated tothe same sense circuit with the same phase may help to mitigateinterference caused by intermodulation effects as described in U.S.patent application Ser. No. 16/392,464 titled Extended Foreign ObjectDetection Signal Processing, the entire contents of which are herebyincorporated by reference.

In some implementations and operations of the multi-purpose detectioncircuit 100, time slots of a scan cycle are reallocated based on someconditions (e.g., whether WPT is active or inactive). In an aspect, itmay be desirable to reduce the detection latency with respect to LODwhen WPT is active. Therefore, in an example operation, two or more timeslots of a scan cycle are allocated to each of the capacitive sensecircuits 108 when WPT is active. Conversely, the LOD function may not berequired when WPT is inactive. Therefore, in an example operation, timeslots of a scan cycle are only allocated to inductive sense circuits(e.g., to the plurality of inductive sense circuits 106) when WPT isinactive. In another example operation, two or more time slots of a scancycle are allocated to each of the plurality of inductive sense circuits(e.g., inductive sense circuits 106) and one time slot is allocated toeach of the plurality of capacitive sense circuits when WPT is inactive.This mode of operation may allow maintaining a limited LOD function whenWPT is inactive (e.g., for purposes of monitoring proper functioning ofthe multi-purpose detection circuit 100 with respect to LOD). Moreover,the time spacing between time slots allocated to the same sense circuitin any of the scanning modes described above is maximized. FIGS. 5A to5E illustrate example implementations of a portion of the multi-purposedetection circuit 100 of FIG. 1 based on inductive sensing by measuringat least one electrical characteristic (e.g., a complex impedance).These examples are to illustrate the principle of the sensing andmeasurement technique and do not show all the details of a multi-purposedetection circuit 100. Particularly, for illustrative purposes, theyonly show a single inductive sense circuit rather than the plurality ofinductive sense circuits (e.g., the plurality of inductive sensecircuits 106 a, 106 b, . . . , 106 n with reference to FIG. 1). Further,they do not show the details of the signal generation, signalprocessing, and evaluation as it may be required e.g., for determiningat least one of a presence of a foreign object, a living object, avehicle, a type of vehicle, and a position of the vehicle and asillustrated by the block diagram of FIG. 4.

The descriptions of the circuits 500, 520, and 540 of FIGS. 5A, 5B, and5C, respectively, are based on measuring a one-port impedance Z₁₁, whilethe circuits 560 and 580 of FIGS. 5D and 5E, respectively, employ atwo-port transimpedance Z₂₁ measurement at the sense frequency e.g.,using a sinusoidal sense signal. However, this should not excludeimplementations configured to measure other electrical characteristicsusing other sense signal waveforms (e.g., multi frequency signals, pulsesignals, pseudo random signals, etc.).

In some implementations, the sense signal is a high frequency signalwith a spectrum substantially in the megahertz (MHz) range (e.g., in afrequency range from 2.5 MHz to 3.5 MHz). In other implementations, thesense signal is constraint to the frequency range from 3.155 MHz to3.400 MHz for frequency regulatory reasons. In some geographic regionsor countries, this frequency range may permit higher emission levelse.g., a magnetic field strength H<13.5 dBμA/m at 10 m from the radiatingparts of the multi-purpose detection circuit 100 (e.g., from theinductive sense element array 107).

The ground symbol shown in the schematic diagrams of FIGS. 5A to 5Eindicate a network node on ground potential referred to as the “circuitground”. However, this should not exclude non-ground-basedimplementations or implementations that use different grounds ondifferent potentials.

The circuit 500 of FIG. 5A illustrates an example implementation basedon measuring a complex impedance Z₁₁ of a one-port inductive sensecircuit 501 (shown in FIG. 5A as the circuit on the right side of thedashed line). More specifically, the impedance Z₁₁ is measured at themeasurement port 508 (indicated in FIG. 5A by a terminal and a dashedline) by applying, from a current source 512 (sense circuit currentsource 512), a sinusoidal current I₀ at the sense frequency with adefined amplitude and phase and by measuring, using a voltagemeasurement circuit 510, the complex open-circuit voltage V (amplitudeand phase) as indicated in FIG. 5A. The impedance Z₁₁ is then determinedby dividing the measured voltage V by the defined (known) current I₀.This impedance measurement technique is also referred herein as to thecurrent source voltage measurement technique.

The sense circuit 501 comprises a single-coil sense element (e.g., sensecoil 502) having an inductance L and an equivalent series resistance R,a series capacitor 504 having a capacitance C_(s) and an equivalentseries resistance R_(Cs) electrically connected in series to the sensecoil 502, and a parallel inductor 506 having an inductance L_(p) and anequivalent series resistance R_(Lp) electrically connected to thecapacitor 504 in parallel to the measurement port 508. The circuit 500further illustrates the sense signal current source 512 and the voltagemeasurement circuit 510 both electrically connected to the sense circuit501 at the measurement port 508.

The equivalent series resistance R includes all electrical lossesintrinsic to the sense coil 502 and extraneous losses as they may occurin its surrounding materials (e.g., the Litz wire of the WPT coil 202and the ferrite of the wireless power transfer structure 200 where thesense coil 502 may be integrated). These materials may interact with themagnetic field as generated by the sense coil 502 causing losses.

The circuit 500 of FIG. 5A also indicates parasitic capacitances (bydashed lines) such as the sense coil's 502 self-capacitance (or intrawinding capacitance) C_(iw), the sense coil's 502 ground capacitanceC_(gnd), and the capacitance C_(wpt) between the sense coil 502 and theWPT coil 202 with reference to FIG. 2 (abstracted in FIG. 5A by a line).These capacitances and the associated electric stray fields may cause acertain sensitivity of the circuit 500 on substantially non-conductive,dielectric objects (e.g., object 112 or 114). For the followingconsiderations, it is assumed that the impact of these capacitances onthe sense coil's 502 impedance is negligible.

The sense circuit 501 may be configured to provide a local minimum inthe impedance magnitude function |Z_(11,0)ω| substantially at a nominalsense frequency, where Z_(11,0) refers to the impedance as presented bythe sense circuit 501 at the measurement port 508 in absence of aforeign object, and ω to the angular frequency. The minimum of theimpedance magnitude is also referred to herein as the series resonanceby definition and applies to the inductive sense circuits with referenceto FIGS. 5A to 5E. Alternatively, the sense circuit 501 may beconfigured to provide a local minimum in the admittance magnitudefunction |Y_(11,0)(ω)| substantially at the nominal sense frequency,where Y_(11,0) (=1/Z_(11,0)) refers to the admittance as presented bythe sense circuit 501 at the measurement port 508 in absence of aforeign object. The minimum of the admittance magnitude is also referredto herein as the parallel resonance by definition and applies to theinductive sense circuits with reference to FIGS. 5A to 5E.

In an example series resonant configuration of the sense circuit 501,the reactance of the series capacitor 504 substantially compensates forthe reactance of the sense coil 502 at the nominal sense frequencyproviding an impedance Z_(11,0) that is substantially real (resistive).In this configuration, the inductance L_(p) of the parallel inductor 506may be similar or larger than the inductance L of the sense coil 502. Inother terms, the impedance magnitude of the parallel inductor 506 may besubstantially (e.g., 10 times) higher than the impedance magnitude|Z_(11,0)| as presented at the nominal sense frequency. In thisconfiguration, the parallel inductor 506 may exert a negligible impacton the impedance |Z_(11,0)| at the nominal sense frequency.

In an example parallel resonant configuration of the sense circuit 501,the reactance of the series capacitor 504 overcompensates for thereactance of the sense coil 502 at the nominal sense frequency. Theresidual capacitive susceptance of the series connection of thecapacitor 504 and the sense coil 502 is substantially compensated for bythe susceptance of the parallel inductor 506 providing an admittanceY_(11,0) that is substantially real (resistive). In this configuration,the inductance L_(p) of the parallel inductor 506 may be smaller,similar, or larger than the inductance L of the sense coil 502. Statedin other terms, the admittance magnitude of the parallel inductor 506may be substantially (e.g., 20 times) higher than the admittancemagnitude |Y_(11,0)| as presented at the nominal sense frequency. Inthis configuration, the parallel inductor 506 exerts a significantimpact on the admittance Y_(11,0) at the nominal sense frequency.

In some implementations, the parallel inductor 506 together with theseries capacitor 504 are used for purposes of resonance tuning andimpedance transformation e.g., to transform the impedance Z₁₁ to matchthe sense circuit 501 with an operating impedance range as previouslymentioned with reference to FIG. 1. The inductance ratio L/L_(p) may bea parameter to control the impedance magnitude |Z_(11,0)|.

Impedance transformation may be particularly effective, if the sensecircuit 501 is configured for parallel resonance. More specifically,increasing the inductance ratio LIL_(p), while maintaining seriesresonance at the nominal sense frequency, may substantially increase theadmittance magnitude |Y_(11,0)| at the nominal sense frequency.Therefore, in an aspect, the sense circuit 501 in the parallel resonantconfiguration may be considered as an alternative to the sense circuit521 illustrated in FIG. 5B using a transformer.

Increasing the inductance ratio L/L_(p), while maintaining resonance atthe nominal sense frequency, may also somewhat decrease the impedancemagnitude |Z_(11,0)| as presented at the nominal sense frequency in theseries resonant configuration of the sense circuit 501. However,impedance transformation may be limited and far less effective than thatof the series resonant configuration.

In another aspect of resonance tuning, the series capacitor 504 mayinclude a variable capacitor whose capacitance C_(s) can beelectronically controlled (e.g., a direct current (DC) controlledcapacitor) forming a variable capacitor 504. In some implementations ofthe circuit 500, a variable capacitor 504 is used to compensate for atemperature drift, an ageing, or a detuning of the sense circuit 701caused by an external impact and to maintain its resonance substantiallyat the nominal sense frequency. Similarly, the parallel inductor 506 mayinclude a variable inductor whose inductance L_(p) can be electronicallycontrolled (e.g., a DC controlled inductor) forming a variable inductor506. In a further aspect, the variable capacitor 504 and variableinductor 506 in combination are used to vary the impedance |Z_(11,0)| ofthe sense circuit 501.

In yet another aspect, the series capacitor 504 in combination with theparallel inductor 506 form a 2^(nd) order high pass filter to attenuatea low frequency disturbance component in the voltage V emanating fromthe voltage inductively coupled into the sense coil 502 by the magneticand electric field as generated during wireless power transfer. Thishigh pass filter may reduce dynamic range requirements of the voltagemeasurement circuit 510 and may also protect the voltage measurementcircuit 510 and the current source 512 from being overloaded. Stated inother words, it may reduce non-linear distortion effects (e.g., signalclipping) in a voltage measurement circuit 510 with a limited dynamicrange.

With reference to FIG. 1, the sense circuit 501, the sense coil 502, theseries capacitor 504, and the parallel inductor 506 may correspond e.g.,to the inductive sense circuit 106 a, the inductive sense element 107 a,and the associated capacitive element, respectively. The current source512 may include the signal generator circuit 406 and the driver circuit402, while the voltage measurement circuit 510 may include themeasurement amplifier circuit 404 and the signal processing circuit 408with reference to FIG. 4.

In an aspect and for sinusoidal signals, a current source (e.g., currentsource 512) may be characterized by a quasi-ideal current sourceproviding a source admittance magnitude |Y_(cs)| substantially (e.g., atleast 10 times) lower than the admittance magnitude |Y₁₁| of the sensecircuit 501 as presented at the measurement port 508 at the sensefrequency. Analogously, the voltage measurement circuit 510 may becharacterized by a quasi-ideal voltage measurement circuit with anadmittance magnitude |Y_(vm)| substantially (e.g., at least 10 times)lower than |Y₁₁| at the sense frequency.

In a further aspect and for sinusoidal signals, a measurement circuit(e.g., measurement circuit 104 of FIG. 4) including a current source(e.g., current source 512) and a voltage measurement circuit (e.g.,voltage measurement circuit 510) configured to measure the admittanceY₁₁ of a one-port sense circuit (e.g., sense circuit 501) may becharacterized by a quasi-ideal measurement circuit providing ameasurement circuit admittance magnitude |Y_(mc)| substantially (e.g.,at least 10 times) lower than |Y₁₁| at the sense frequency, where themeasurement circuit admittance may be defined, using above admittancedefinitions, as:

Y _(mc) ≈Y _(cs) +Y _(vm)  (1)

Conversely, the quality of a measurement circuit (e.g., measurementcircuit 104 of FIG. 4) based on the current source voltage measurementapproach may be characterized as the ratio:

Q _(mc) ≈|Y ₁₁ |/|Y _(mc)|  (2)

Equation (2) may be used to assess the quality of a measurement circuit(e.g., measurement circuit 104 of FIG. 4) based on the current sourcevoltage measurement approach.

A more general definition of the quality of a measurement circuit (e.g.,measurement circuit 104 of FIG. 4) based on the current source voltagemeasurement approach, also applicable to a two-port sense circuit (e.g.,sense circuit 561 with reference to FIG. 5D) may be given by:

Q _(mc) ≈|ΔV/V ₀ |/|ΔI/I ₀|  (3)

Above characterizations of the current source 512, the voltagemeasurement circuit 510, and the measurement circuit 104 may begeneralized to non-sinusoidal sense signals, where the notions ofcomplex impedance and complex amplitude may not directly apply. This maybe accomplished by approximating the signal by a complex Fourier seriesand applying above characterizations to the individual frequencycomponents of the complex Fourier series.

Other impedance measurement techniques may also be contemplated e.g., byapplying a sinusoidal voltage, from a voltage source (e.g., voltagesource 552 with reference to FIG. 5C) with a defined voltage V₀(amplitude and phase) to the sense circuit 501 and by measuring thecomplex current I (amplitude and phase) at the measurement port 508using a current measurement circuit (e.g., current measurement circuit550 with reference to FIG. 5C).

Analogously to the current source voltage measurement technique, thevoltage source 552 (sense signal voltage source 552) may becharacterized by a quasi-ideal voltage source with a source impedancemagnitude |Z_(vs)| substantially (e.g., at least 10 times) lower thanthe impedance magnitude |Z₁₁| of the sense circuit 501 as presented atthe sense frequency. Analogously, the current measurement circuit 550may be characterized by a quasi-ideal current measurement circuit withan impedance magnitude |Z_(cm)| substantially (e.g., at least 10 times)lower than |Z₁₁| at the sense frequency.

In a further aspect, a measurement circuit (e.g., measurement circuit104 of FIG. 4) including a voltage source (e.g., voltage source 552) anda current measurement circuit (e.g., current measurement circuit 550)may be characterized by a quasi-ideal measurement circuit providing ameasurement circuit impedance magnitude |Z_(mc)| substantially (e.g., atleast 10 times) lower than |Z₁₁| at the sense frequency, where themeasurement circuit impedance may be defined, using above impedancedefinitions, as:

Z _(mc) ≈Z _(vs) +Z _(cm)  (4)

Conversely, the quality of a measurement circuit (e.g., measurementcircuit 104 of FIG. 4) based on the voltage source current measurementapproach may be characterized as the ratio:

Q _(mc) ≈|Z ₁₁ |/|Z _(mc)|  (5)

Equation (5) may be used to assess the quality of a measurement circuit(e.g., measurement circuit 104 of FIG. 4) based on the voltage sourcecurrent measurement approach.

Other impedance measurement techniques may also include approaches wherethe sense circuit 501 is driven by a non-ideal source and the voltage Vand the current I are measured e.g., using a quasi-ideal voltagemeasurement circuit and a quasi-ideal current measurement circuit,respectively.

Further, in some implementations, measurement of the voltage V and thusof the impedance Z₁₁ may be affected by noise and other disturbancesignals reducing a detection sensitivity of the multi-purpose detectioncircuit 100. The noise may include circuit intrinsic noise as generatedin active and passive components of the circuit 500 of FIG. 5A. It mayalso include quantization noise e.g., generated in a digitalimplementation of the signal generator circuit 406 and the signalprocessing circuit 408 with reference to FIG. 4. Other disturbancesignals may emanate from sources external to the circuit 500 (e.g., fromthe WPT system during wireless power transfer, from a switched-modepower supply, from a digital processing unit, etc.). These circuitextrinsic disturbance signals may be inductively and capacitivelycoupled (e.g., via capacitance C_(wpt)) to the sense coil 502 and mayinclude the fundamental and harmonics of the WPT operating frequency andother switching noise components as generated by the WPT system.Therefore, in some implementations, the voltage measurement circuit 510includes a filter to selectively filter the sense signal and to suppressnoise and other disturbance signal components as discussed above andconsequently to improve the detection sensitivity. The filter may bematched to the sense signal and configured to maximize a signal-to-noiseratio (SNR) in presence of noise and other disturbance signals. Inimplementations using a sinusoidal sense signal, the voltage measurementcircuit 510 may be frequency selective (narrowband) and tuned to thesense signal frequency. It may be configured to suppress noise and otherdisturbance signal components at frequencies substantially differentfrom the sense frequency.

Moreover, in implementations employing a selective voltage measurementcircuit 510 as discussed above, the sense signal waveform as generatedby the current source 512 and the corresponding filter of the voltagemeasurement circuit 510 are adapted e.g., to improve the SNR andconsequently to improve the detection sensitivity. Therefore, in someimplementations, the voltage measurement circuit 510 also includes anoise analyzer (e.g., included in the signal processing circuit 408 withreference to FIG. 4) that is continuously analyzing the noise. Further,it includes a controller (e.g., the control and evaluation circuit 102of FIG. 4) for controlling the waveform of the sense signal as generatedby the current source 512 based on the noise analysis and within someoperational constraints. More specifically, in an example implementationusing a sinusoidal sense signal, the voltage measurement circuit 510includes a spectrum analyzer and a controller that is continuouslylooking for frequencies with a minimum disturbance (noise) level andadjusts the frequency of the sense signal (sense frequency) to afrequency with the minimum disturbance level, avoiding switchingharmonics of the WPT system and remaining substantially at resonance ofthe sense circuit 501.

With reference to FIG. 1, FIG. 5A also illustrates objects 110, 112, and114 proximate to the sense coil 502. Presence of any one of theseobjects including vehicle 330 may cause a change in one or moreelectrical characteristics of the sense coil 502 and consequently of thesense circuit 501. As non-limiting examples, it may cause a change in atleast one of the inductance L and the equivalent series resistance R andhence in the sense coil's 502 impedance Z. This change of impedance,herein referred to as the reflected impedance ΔZ _(r) of the object(e.g., object 110), results in an impedance change ΔZ with respect tothe impedance Z_(11,0) as presented at the measurement port 508 inabsence of a foreign object. As discussed below in more detail withreference to FIG. 6 and FIG. 5F, the reflected impedance ΔZ _(r) and therelated impedance change ΔZ may be indicative of electrical propertiesof the object (e.g., object 110).

Presence of an object (e.g., object 110) may be determined if ΔZsatisfies certain criteria (e.g., magnitude |ΔZ| exceeding a detectionthreshold, angle arg{ΔZ} being within a certain range). Though not shownin FIG. 5A, a change ΔZ _(r) in Z and thus ΔZ in the impedance Z₁₁ mayalso be caused by the underbody of a vehicle (e.g., vehicle 330), by thevehicle-based wireless power transfer structure (e.g., wireless powertransfer structure 310 of FIG. 3), by a passive beacon transponder(e.g., passive beacon transponder 314 of FIG. 3), or by anotherstructure at the vehicle. Therefore, a change ΔZ may be also indicativeof the presence of a vehicle or a type of vehicle above the sense coil502. Further, an impedance change ΔZ may be caused by a substantiallynon-conductive, dielectric object (e.g., object 112 or 114) proximate tothe sense coil 502 due to the capacitive sensing effect inherent to thesense coil 502 as previously mentioned. More specifically, the object112 or 114 in proximity of the sense coil 502 may change one or more ofits parasitic capacitances C_(iw), C_(gnd), and C_(wpt) as illustratedin FIG. 5A.

In an implementation of the circuit 500 based on measuring theadmittance Y₁₁, presence of an object (e.g., object 110, 112, 114, orvehicle 330) may cause a change ΔY with respect to the admittanceY_(11,0) as measured in absence of a foreign object. Analogously,presence of an object (e.g., object 110) may be determined if ΔYsatisfies certain criteria (e.g., magnitude ILYI exceeding a detectionthreshold, angle arg{ΔY}) being within a certain range).

Using a quasi-ideal current source (e.g., the current source 512), achange ΔZ in the impedance Z₁₁ (e.g., due to the presence of the object110) manifests in a change ΔV in the voltage V while the current I₀remains substantially unaffected. Therefore, measuring the complexvoltage V may be equivalent to measuring the complex impedance Z₁₁. Inother words, the complex voltage V may be indicative of the compleximpedance Z₁₁ and there may be no requirement for additionally measuringthe current I₀ thus reducing complexity of the measurement circuit(e.g., measurement circuit 104 of FIG. 1). Likewise, measuring thecomplex voltage V and determining the reciprocal value 1/V may beequivalent to measuring the complex admittance Y₁₁.

In an aspect, it may be useful to define the normalized reflectedimpedance of an object (e.g., object 110) in the sense coil's 502 havinga reactance ω L as:

ΔZ _(r)′=(Z−jωL)/(ωL)=ΔZ _(r)/(ωL)  (6)

where Z defines the sense coil's 502 impedance in presence of an object(e.g., object 110). Analogously, the normalized reflected admittanceΔY_(r)′ may be defined as:

ΔY _(r)′=(Y−(1/(jωL))ωL=ΔY_(r)ωL  (7)

where Y and ΔY_(r) denote the sense coil's 502 admittance in presence ofan object (e.g., object 110) and the reflected admittance of the object,respectively. The normalized reflected impedance ΔZ_(r)′ or thenormalized reflected admittance ΔY_(r)′ determine the impact of anobject (e.g., object 110) on the sense coil's 502 impedance oradmittance, respectively. Its magnitude |ΔZ_(r)′| or |ΔY_(r)′| may berelated to the size, the position, and orientation of the objectrelative to the sense coil 502.

In a further aspect, it may be useful to define the normalized impedancechange of a one-port sense circuit (e.g., sense circuit 501 of FIG. 5A)as:

ΔZ′=(Z ₁₁ −Z _(11,0))/|Z _(11,0) |=ΔZ/|Z _(11,0)|  (8)

and analogously, the normalized admittance change as:

ΔY′=(Y ₁₁ −Y _(11,0)/|Y _(11,0) |=ΔY/|Y _(11,0)|  (9)

also referred herein as to the fractional change ΔZ′ (or ΔY′). Thefractional change ΔZ′ (or ΔY′) caused by a defined test object (e.g.,object 110) placed at a defined position relative to the sense coil 502may relate to the detection sensitivity of an object detection circuit(e.g., the multi-purpose detection circuit 100 of FIG. 1) based on theone-port sense circuit 501. More specifically, increasing the fractionalchange ΔZ′ (or ΔY′) may increase a signal-to-noise ratio (SNR) e.g.,defined as:

ΔSNR=|ΔV|/V _(n)  (10)

with V_(n) (not indicated in FIG. 5A) referring to the noise componentin the voltage V. In another aspect, increasing the fractional changemay reduce dynamic range requirements of the voltage measurement circuit510.

As non-limiting examples, the normalized reflected impedance ΔZ_(r)′ ofan object (e.g., object 110) and thus the related fractional change ΔZ′may be increased by optimizing the design of the sense coil 502 withrespect to its geometry and its integration into the wireless powertransfer structure (e.g., wireless power transfer structure 200 withreference to FIGS. 2 and 3). The fractional change ΔZ′ may be furtherincreased by resonance tuning e.g., using the series capacitor 504, andby improving the Q-factor of the sense circuit 501. Improving theQ-factor may also increase the SNR, if the noise voltage V_(n) ispredominantly circuit intrinsic noise as discussed below with referenceto FIG. 5F. The same may apply to the normalized reflected admittanceΔY_(r)′ and the fractional change ΔY′.

As further analyzed and discussed below with reference to FIG. 5F, useof the parallel inductor 506 for purposes of parallel resonance tuningand impedance transformation may result in a lower fractional change ascompared to the sense circuit 501 using the parallel inductor 506 onlyfor purposes of high pass filtering as previously discussed. This may beexplained by the additional losses inherent to the parallel inductor506.

In a further aspect of the multi-purpose detection circuit 100,variations in temperature e.g., of the sense coil 502 may result inthermal drift of the impedance Z₁₁ as measured at the measurement port508. In some implementations, the sense coil's inductance L andequivalent series resistance R, the series capacitor's 504 capacitance,and the parallel inductor's 506 inductance L_(p) and equivalent seriesresistance R_(Lp) may be subjected to thermal drift. Thermal drifteffects may deteriorate the detection sensitivity of the multi-purposeobject detection circuit 100. Considering the physical nature oftemperature drifts in a tuned sense circuit (e.g., sense circuit 501),it may be meaningful to define a temperature sensitivity S_(ϑ) for thereal and imaginary part, separately, as the ratios:

Re{S _(ϑ) }=Re{ΔZ _(ϑ) ′}/Re{ΔZ′}  (11)

Im{S _(ϑ) }=Im{ΔZ _(ϑ) ′}/Im{ΔZ′}  (12)

where ΔZ_(ϑ)′ denotes the fractional impedance change due to a definedtemperature change Δϑ and ΔZ′ the fractional impedance change due topresence of a test object (e.g., object 110) at a defined positionrelative to the sense coil 502. The fractional change ΔZ_(ϑ)′ may beconsidered the complex temperature coefficient of a sense circuit (e.g.,sense circuit 501). The temperature sensitivity S_(ϑ) may also beexpressed in terms of the fractional admittance changes ΔY_(ϑ)′ and ΔY′.

In yet another aspect of the multi-purpose detection circuit 100, it maybe desirable to discriminate between certain categories of objects e.g.,between foreign metallic objects (e.g., object 110), non-livingnon-conductive objects (e.g., object 112), and living objects (e.g.,object 114). In another aspect, it may also be desirable to discriminatee.g., between foreign metallic objects (e.g., object 110) and thevehicle 330 with reference to FIG. 3. As further discussed below withreference to FIG. 6, this may be accomplished based on characteristicsof the reflected impedance ΔZ_(r) as defined above. As already mentionedabove and discussed in more details with reference to FIG. 6, thereflected impedance ΔZ_(r) and particularly the angle arg{ΔZ_(r)} mayreflect electrical properties of the object 110, 112, 114, or vehicle330. The same may be true for the reflected admittance ΔY_(r).

In some implementations and configurations of the circuit 500 of FIG.5A, the change ΔZ in the impedance Z₁₁ caused by an object (e.g., object110) is indicative of the reflected impedance ΔZ_(r). Therefore, in anaspect of object discrimination, the circuit 500 may be configured todetermine the angle arg{ΔZ} with the required accuracy. However, in someimplementations, measuring the angle arg{ΔZ} may be subject to errorsfor various reasons. One prominent error source of some implementationsof the circuit 500 is an unknown (e.g., frequency dependent) phaseoffset of the output of the voltage measurement circuit 510 relative tothe drive current I₀ as generated by the current source 512. In a mixeddigital and analog implementation of the circuit 500, this phase offsetmay be attributed to the analog frontend portion of the circuit 500.

In an aspect of reducing an error in the measurement of the anglearg{ΔZ}, some implementations of a multipurpose detection circuit 100employ a phase calibration of the analog circuitry (e.g., the analogfront end portion of the measurement circuit 104 with reference to FIG.4). This phase calibration may be a factory calibration or it may beperformed at the time of installation and commissioning of the wirelesspower transfer structure 200 (integrating the multipurpose detectioncircuit 100). In some operations of the multipurpose detection circuit100, this phase calibration is repeated periodically in fixed intervals(e.g., to mitigate ageing effects). In other operations, it is executedafter the multipurpose detection circuit 100 is reactivated (poweredon). In further operations, this calibration is initiated e.g., if thetemperature as measured in the wireless power transfer structure 200exceeds or falls below a threshold.

Reactance compensation (resonance tuning) in the sense circuit 501produces a local extremum (minimum or maximum) in the impedancemagnitude function |Z_(11,0)(ω)| and hence in the voltage magnitude |V|across the measurement port 508. Therefore, reactance compensationprovides a mean to calibrate the voltage measurement circuit 510 andhence the impedance measurement with respect to the angle arg{Δz}.

In a first step of an example calibration procedure applicable to theseries resonant configuration of the circuit 500 of FIG. 5A, the sensefrequency is adjusted to the local minimum of the voltage magnitude |V|as measured by the voltage measurement circuit 510 supposing absence ofa foreign object. At this frequency, the complex impedance Z_(11,0) andhence the complex voltage V across the measurement port 508 may besubstantially real. Otherwise stated, the angles arg{Z_(11,0)} andarg{V} are substantially zero. In a second step of the examplecalibration procedure, the voltage measurement circuit 510 is correctedby applying a phase shift such that the imaginary part of the complexvoltage value as determined and output by the voltage measurementcircuit 510 at this frequency vanishes. Applying the phase shift isequivalent to rotating the impedance plane by an angle arg{V_(uncal)}where V_(uncal) refers to the complex voltage value as determined by theuncalibrated voltage measurement circuit 510 (before any correction isapplied). This angle correction may be expressed by the followingcomplex multiplication:

V _(cal) =V _(uncal) exp(−j arg{V _(uncal)})  (13)

where V_(cal) refers to the complex voltage value as determined by thecalibrated voltage measurement circuit 510.

Applying the angle correction of Equation (13), an object (e.g., object110) reflecting an impedance ΔZ_(r) that is imaginary (reactive) maycause a measured voltage change ΔV_(cal) that is substantiallyimaginary. Nevertheless, a small residual error may remain in the anglearg{ΔV_(cal)} due to the impact of the parallel inductor 506 and theelectrical losses in the sense circuit 501. The residual angle error ofan example series resonant configuration of the circuit 500 and for anexample object 110 is provided in TABLE 2.

In some implementations, the residual error described above is reducedby configuring the parallel inductor 506 with an inductance L_(p) whoseimpedance Z_(Lp) is substantially larger (e.g., 10 times larger) thanthe series resonant resistance of the sense circuit 501. In otherimplementations, the residual error is reduced by measuring theimpedance Z_(11,0) at two or more substantially different frequenciesand by determining the elements of an equivalent circuit model of thesense circuit 501 (e.g., the equivalent circuit model illustrated inFIG. 5F) based on the measured impedances Z_(11,0) employing a best fitmethod. In some implementations, these two or more frequencies includeat least the frequency of the minimum and the maximum of |Z_(11,0)(ω)|.

In an implementation of the multipurpose detection circuit 100 using aplurality of inductive sense circuits (e.g., inductive sense circuits106 a, 106 b, . . . , 106 n), each including a respective inductivesense element (e.g., inductive sense element 107 a, 107 b, . . . , 107n) of an array (e.g., array 107), a further residual error may be causedby a parasitic resonance effect of sense circuits associated to adjacentinductive sense elements. More precisely, a residual error in a firstsense circuit (e.g., inductive sense circuit 106 a) including a firstinductive sense element (e.g., inductive sense element 107 a) may becaused by a parasitic resonance effect of at least one second inductivesense circuit (e.g., inductive sense circuit 106 b) including a secondinductive sense element (e.g., inductive sense element 107 b) that islocated adjacent to the first inductive sense element.

Therefore, in some implementations of the multipurpose detection circuit100, the measurement accuracy of the angle arg{ΔZ} and thus of the anglearg{ΔZ_(r)} is increased by an optimized design of the sense coil 502and by introducing some spacing between adjacent sense coils 502 of anarray (e.g., array 107).

In an implementation configured for parallel resonance as defined above,the circuit 500 may be configured to measure the admittance Y₁₁ andcorresponding changes ΔY of Y₁₁ as caused by the object 110, 112, 114,or vehicle 330. In this case, the admittance change ΔY may be indicativeof the reflected impedance ΔZ_(r) as previously introduced. As discussedabove with reference to the series resonant configuration, the anglearg{ΔY} may be subjected to an error and therefore may requirecalibration to reduce an error in the measurement of the angle arg{ΔY}and thus of the angle arg{ΔZ_(r)}.

In an implementation configured for parallel resonance, the circuit 500may be calibrated analogously to the series resonant configuration usingthe local minimum of the admittance function |Y_(11,0)(ω)| wheresusceptance compensation occurs.

In a first step of an example calibration procedure applicable to theparallel resonant configuration of the circuit 500 of FIG. 5A, the sensefrequency is adjusted to the local maximum of the voltage magnitude |V|as measured by the uncalibrated voltage measurement circuit 510supposing absence of a foreign object. At this frequency, the admittanceY_(11,0) and hence the voltage V across the measurement port 508 may besubstantially real. Otherwise stated, the angles arg{Y_(11,0)} andarg{V} are substantially zero. In a second step of the examplecalibration procedure, the voltage measurement circuit 510 is correctedby applying a phase shift (impedance plane rotation) as defined above byEquation (13).

Applying the angle correction of Equation (13), an object (e.g., object110) reflecting an impedance ΔZ_(r) that is imaginary (reactive) mayresult in a measured voltage change ΔV_(cal) that is substantiallyimaginary. A residual error may remain in the angle arg{ΔV_(cal)} due tothe transformation of ΔZ_(r) to ΔY in the lossy sense circuit 501. Theresidual angle error of an example parallel resonant configuration ofthe circuit 500 and for example reflected impedance ΔZ_(r) is providedin TABLE 2.

In an example implementation, the residual error due to thetransformation of ΔZ_(r) to ΔY is reduced by measuring the admittanceY_(11,0) at two or more substantially different frequencies, supposingabsence of a foreign object, and by determining the elements of anequivalent circuit model (e.g., the equivalent circuit model of FIG. 5F)based on the measured admittances Y_(11,0) employing a best fit method.In some implementations, these two or more frequencies include at leastthe frequency of the minimum and the maximum of |Y_(11,0)(ω)|.

The series and the parallel resonant configuration of the circuit 500 ofFIG. 5A are further analyzed below with reference to FIG. 5F withrespect to various characteristics such as the Q-factor, fractionalchange, and various definitions of SNR based on an equivalent circuitmodel.

The circuit 520 of FIG. 5B illustrates another example implementationbased on measuring a complex impedance Z₁₁ of a one-port inductive sensecircuit 521 (shown in FIG. 5B as the circuit on the right side of thedashed line). More specifically, the impedance Z₁₁ is measured at themeasurement port 528 (indicated in FIG. 5B by a terminal and a dashedline) by applying, from the current source 512, a sinusoidal current I₀and by measuring, using the voltage measurement circuit 510, the complexopen-circuit voltage V as previously described with reference to FIG.5A.

The sense circuit 521 comprises the single-coil inductive sense element(e.g., sense coil 502) having the inductance L with reference to FIG. 5Aand a capacitor 524 having a capacitance C_(s) electrically connected inseries to the sense coil 502. However, the sense circuit 521 shows theparallel inductor 506 of FIG. 5A replaced by a transformer 526. Thetransformer 526 may include a primary winding and a galvanicallyinsulated secondary winding wound on a common core as suggested by thetransformer symbol in FIG. 5B. However, other transformerimplementations may apply e.g., an autotransformer having only onewinding with at least three terminals. FIG. 5B also indicates atransformation ratio n_(T):1, a main inductance L_(m), a leakageinductance L_(σ), and equivalent series resistances R_(Lm) and R_(w)that may represent core and conductor losses, respectively. Theseparameters may refer to the secondary referred approximate equivalentcircuit model of a non-ideal transformer illustrated in FIG. 5H. FIG. 5Bshows its primary winding electrically connected in parallel to themeasurement port 528, while its secondary winding is electricallyconnected to the series capacitor 524. The circuit 520 furtherillustrates the sense signal current source 512 and the voltagemeasurement circuit 510 both electrically connected to the sense circuit521 at the measurement port 528.

Though not indicated in FIG. 5B for purposes of illustration, the seriescapacitor 524 and the sense coil 502 may also include the equivalentseries resistance R_(Cs) and the parasitic capacitances C_(iw), C_(gnd),and C_(wpt), respectively, as shown in FIG. 5A.

The sense circuit 521 may be configured to provide a local minimum inthe impedance magnitude function |Z_(11,0)(ω)| (series resonance)substantially at the nominal sense frequency. Alternatively, it may beconfigured to provide a local minimum in the admittance magnitudefunction |Y_(11,0)(ω)| (parallel resonance) substantially at the nominalsense frequency using the transformer's 526 secondary referred maininductance L_(m) in a manner similar to using the inductance L_(p) asdescribed above with reference to FIG. 5A.

In an example series resonant configuration of the sense circuit 521,the reactance of the series capacitor 504 substantially compensates forthe reactance of the sense coil 502 at the nominal sense frequencyproviding an impedance Z_(u)o at the measurement port 528 that issubstantially real (resistive). The reactance of the series capacitor524 also compensates for the reactance of the transformer's 526secondary referred leakage inductance L_(σ) with reference to FIG. 5H.In this configuration, the transformer's 526 secondary referred maininductance L_(m) may be similar or larger than the inductance L of thesense coil 502. Stated in other terms, the primary referred open-circuitimpedance of the transformer 526 may be substantially (e.g., 10 times)higher than the impedance magnitude |Z_(11,0)| as presented at thenominal sense frequency. Apart from the impedance transformation by thefactor n_(T) ², the transformer 526 may exert a negligible impact on theimpedance |Z_(11,0)| at the nominal sense frequency.

In an example parallel resonant configuration of the sense circuit 521,the reactance of the series capacitor 524 overcompensates for the sumreactance of the sense coil 502 and the transformer's 526 leakageinductance L_(σ) at the nominal sense frequency. The residual capacitivesusceptance of the series connection of the capacitor 524, the sensecoil 502 and the transformer's leakage inductance L_(σ) is substantiallycompensated for by the susceptance of the transformer's 526 secondaryreferred inductance L_(m) providing an admittance Y_(11,0) that issubstantially real (resistive). In this configuration, the inductanceL_(m) may be smaller, similar, or larger than the inductance L of thesense coil 502. Stated in other terms, the primary referred open-circuitadmittance of the transformer 526 may be substantially (e.g., 20 times)higher than the admittance magnitude |Y_(11,0)| as presented at thenominal sense frequency. In this configuration and apart from theadmittance transformation, the transformer 526 exerts a significantimpact on the admittance Y_(11,0) at the nominal sense frequency.

The transformer 526 may serve for various purposes. In someimplementations, the transformer 526 is a n_(T):1 transformer withn_(T)≠1 used at least for impedance transformation e.g., to match theimpedance magnitude |Z₁₁| of the sense circuit 521 with an operatingimpedance range as previously mentioned with reference to FIG. 5A. In anexample implementation configured for series resonance, the transformer526 increases the impedance |Z₁₁| by a factor n_(T) ² with n_(T)>1. Inanother example implementation configured for parallel resonance, itincreases the admittance |Y₁₁| by a factor 1/n_(T) ² with n_(T)<1. Inyet other implementations, it is a balancing (balun) transformer used toreduce a common mode disturbance voltage capacitively coupled to thesense coil 502 (e.g., via parasitic capacitance C_(wpt)). In a furtherimplementation, it is a balancing transformer used to reduce a groundleakage current e.g., via parasitic capacitance C_(gnd) and thus toreduce at least one of a sensitivity to living objects (e.g., livingobject 112) and an electromagnetic emission. In yet anotherimplementation, the transformer 526 is also part of the resonance tuningas described above.

Apart from the transformation ratio n_(T):1, the inductance ratioL/L_(m) may be an additional parameter to match the admittance magnitude|Y_(11,0)| of the parallel resonant configuration with an operatingadmittance range of the multi-purpose object detection circuit 100 in amanner similar to the parameter L/L_(p) in the circuit 500 of FIG. 5A.

In a further aspect, the series capacitor 524 in combination with thetransformer's 526 main inductance L_(m) form a 2^(nd) order high passfilter to attenuate a low frequency disturbance component in the voltageV for purposes as previously discussed in connection with FIG. 5A.

FIG. 5B also illustrates the objects 110, 112, and 114 (with referenceto FIG. 1) proximate to the sense coil 502. As previously discussed withreference to FIG. 1, presence of an object (e.g., object 110, 112, 114,or vehicle 330) may cause a change in one or more electricalcharacteristics of the sense coil 502 and consequently of the sensecircuit 521. Not limited to that, the object may change the sense coil'simpedance Z referred to as the reflected impedance ΔZ_(r) with referenceto FIG. 5A.

The losses of the transformer 526 and its leakage inductance L_(σ) maysomewhat reduce the fractional change ΔZ (or ΔY) of the sense circuit521 if compared to the transformerless sense circuit 501 of FIG. 5A.This is further analyzed and discussed below with reference to FIG. 5F.

The circuit 540 of FIG. 5C illustrates another example implementationbased on measuring a complex impedance Z₁₁ of a one-port inductive sensecircuit 541 (shown in FIG. 5C as the circuit on the right side of thedashed line). More specifically, the impedance Z₁₁ is measured at themeasurement port 548 (indicated in FIG. 5C by a terminal and a dashedline) by applying, from a voltage source 552, a sinusoidal voltage V₀and by measuring, using a current measurement circuit 550, the complexshort-circuit current I as previously mentioned with reference to FIG.5A (voltage source current measurement technique).

The circuit 540 may be considered an electrically dual circuit of thecircuit 500 of FIG. 5A according to the principle of duality inelectrical engineering. The circuit 540 includes the sense circuit 541comprising the sense coil 502 having an inductance L with reference toFIG. A, a parallel capacitor 544 having a capacitance C_(p) electricallyconnected in parallel to the sense coil 502, and a series capacitor 546having a capacitance C_(s) electrically connected in series to theparallel connection of the sense coil 502 and parallel capacitor 544.The circuit 540 further illustrates the sense signal voltage source 552and the current measurement circuit 550 both electrically connected tothe sense circuit 541 at the measurement port 548.

In another aspect, the sense circuit 541 may also include a transformer(not shown herein) e.g., electrically connected between the measurementport 548 and the capacitor 546 e.g., for purposes of balancing.

Though not indicated in FIG. 5C for purposes of illustration, thecapacitive and inductive elements of the sense circuit 541 may alsocause electrical losses that may be represented by a respectiveequivalent series resistance as previously discussed with reference toFIG. 5A. Further, the sense coil 502 may also include the parasiticcapacitances C_(iw), C_(gnd), and C_(wpt) as indicated in FIG. 5A bydashed lines.

As with the circuit 500 of FIG. 5A, the circuit 540 of FIG. 5C may beconfigured to provide a local minimum in the admittance magnitudefunction |Y_(11,0)(ω)|) substantially at the nominal sense frequency.Alternatively, it may be configured to provide a series resonance (alocal minimum in the impedance magnitude function|Z_(11,0)(ω)|=1/|Y_(11,0)(ω)|) substantially at the nominal sensefrequency.

In an example parallel resonant configuration of the sense circuit 541,the susceptance of the parallel capacitor 544 substantially compensatesfor the susceptance of the sense coil 502 at the nominal sense frequencyproviding an admittance Y_(11,0) that is substantially real (resistive).In this configuration, the capacitance C_(s) of the series capacitor 546may be similar or larger than the capacitance C_(p) of the parallelcapacitor 544. Stated in other terms, the admittance magnitude of theseries capacitor 546 may be substantially (e.g., 10 times) higher thanthe admittance magnitude |Y_(11,0)| as presented at the nominal sensefrequency. In this configuration, the series capacitor 546 may exert anegligible impact on the admittance |Y_(11,0)| at the nominal sensefrequency.

In an example series resonant configuration of the sense circuit 541,the susceptance of the parallel capacitor 544 undercompensates for thesusceptance of the sense coil 502 at the nominal sense frequency. Theresidual inductive reactance of the parallel connection of the capacitor544 and the sense coil 502 is substantially compensated for by thereactance of the series capacitor 546 providing an impedance |Z_(11,0)|that is substantially real (resistive). In this configuration, thecapacitance C_(s) of the series capacitor 546 may be smaller, similar,or larger than the capacitance C_(p) of the parallel capacitor 544.Stated in other terms, the impedance magnitude of the series capacitor546 may be substantially (e.g., 20 times) higher than the impedancemagnitude |Z_(11,0)| as presented at the nominal sense frequency. Inthis configuration, the series capacitor 546 exerts a significant impacton the impedance Z_(11,0) at the nominal sense frequency.

In some implementations, the series capacitor 546 together with theparallel capacitor 544 are used for purposes of resonance tuning andimpedance transformation e.g., to transform the impedance Z₁₁ to matchthe sense circuit 541 with an operating impedance range as previouslymentioned with reference to FIG. 1. The capacitance ratio C_(p)/C_(s)may be a parameter to control the impedance magnitude |Z_(11,0)|.

Impedance transformation may be particularly effective, if the sensecircuit 541 is configured for series resonance. More specifically,increasing the capacitance ratio C_(p)/C_(s), while maintaining seriesresonance at the nominal sense frequency, may substantially increase theimpedance magnitude |Z_(11,0)| at the nominal sense frequency.Therefore, in an aspect, the sense circuit 541 in the series resonantconfiguration may be considered as an alternative to the sense circuit521 of FIG. 5B using the transformer 726.

Increasing the capacitance ratio C_(p)/C_(s), while maintainingresonance at the nominal sense frequency, may also somewhat decrease theadmittance magnitude |Y_(11,0)| as presented at the nominal sensefrequency in the parallel resonant configuration of the sense circuit541. However, impedance transformation may be limited and far lesseffective than that of the series resonant configuration.

In a further aspect, the sense circuit 541 due to the series capacitor546 in conjunction with the voltage source current measurement techniqueprovides a high pass filter characteristic to attenuate a low frequencydisturbance component in the current/emanating from the voltageinductively coupled into the sense coil 502 by the magnetic and electricfield as generated during wireless power transfer. This high pass filtermay reduce dynamic range requirements of the current measurement circuit550 and may also protect the current measurement circuit 550 and thevoltage source 552 from being overloaded. Stated in other terms, it mayreduce non-linear distortion effects (e.g., signal clipping) in acurrent measurement circuit 550 with a limited dynamic range.

With reference to FIG. 1, the sense circuit 541, the sense coil 502, theparallel capacitor 544, and the series capacitor 546 may corresponde.g., to the inductive sense circuit 106 a, the inductive sense element107 a, and the associated capacitive element, respectively. The voltagesource 552 may include the signal generator circuit 406 and the drivercircuit 402, while the current measurement circuit 550 may include themeasurement amplifier circuit 404 and the signal processing circuit 408with reference to FIG. 4.

In some implementations, the voltage source 552 may be characterized bya quasi-ideal voltage source providing a source impedance whosemagnitude is substantially (e.g., 10 times) lower than the magnitude ofthe impedance |Z₁₁| of the sense circuit 541 as presented at the sensefrequency. Analogously, the current measurement circuit 550 may becharacterized by a quasi-ideal current measurement circuit with animpedance magnitude substantially (e.g., 10 times) lower than theimpedance magnitude |Z₁₁| at the sense frequency.

Above characterizations of the voltage source 552 and the currentmeasurement circuit 550 may be generalized to non-sinusoidal sensesignals as previously discussed with reference to FIG. 5A.

Other impedance measurement techniques may also be contemplated e.g., byapplying a sinusoidal current, from the current source 512, with adefined current I₀ (amplitude and phase) to the sense circuit 541 and bymeasuring the complex voltage V (amplitude and phase) at the measurementport 548 using the voltage measurement circuit 510 as previouslydiscussed with reference to FIG. 5A.

Further, in some implementations, measurement of the current I and thusof the impedance Z₁₁ may be affected by noise and other disturbancesignals reducing a detection sensitivity of the multi-purpose detectioncircuit 100 as previously discussed with reference to FIG. 5A.Therefore, in some implementations, the current measurement circuit 550includes a filter to selectively filter the sense signal and to suppressnoise and other disturbance signal components and consequently toimprove the detection sensitivity as previously discussed.

With reference to FIG. 1, FIG. 5C also illustrates the objects 110, 112,and 114 proximate to the sense coil 502. Presence of the object 110,112, 114, or vehicle 330 (not shown in FIG. 5C) may cause a change inone or more electrical characteristics of the sense coil 502 andconsequently of the sense circuit 541. As non-limiting examples, it maycause a change of the sense coil's 502 admittance Y referred to as thereflected admittance ΔY_(r) with reference to FIG. 5A, a change ΔY withrespect to the admittance Y_(11,0) as measured in absence of a foreignobject.

Presence of an object (e.g., object 110) may be determined if ΔYsatisfies certain criteria (e.g., magnitude |ΔY| exceeding a detectionthreshold, angle arg{ΔY} being within a certain range). In animplementation of the circuit 540 where the impedance Z₁₁ is measured aspreviously mentioned in connection with the series resonance, presenceof an object (e.g., object 110) may cause a change ΔZ with respect tothe impedance Z_(11,0).

Using a quasi-ideal voltage source 552, a change ΔY in the admittanceY₁₁ (e.g., due to presence of the object 110) manifests in a change ΔIin the current I while the voltage V₀ remains substantially unaffected.Therefore, measuring the complex current I may be equivalent tomeasuring the complex admittance Y₁₁. In other words, the complexcurrent I may be indicative of the complex admittance Y₁₁ and there maybe no requirement for additionally measuring the voltage V₀ thusreducing complexity of the measurement circuit (e.g., measurementcircuit 104 of FIG. 1)

The fractional change ΔY′ (or ΔZ′) as defined by Equations (8) and (9)and with respect to a defined test object (e.g., object 110) placed at adefined position relative to the sense coil 502 may relate to thedetection sensitivity of an object detection circuit (e.g., themulti-purpose detection circuit 100 of FIG. 1) based on the sensecircuit 541. More specifically, increasing the fractional change ΔY′ (orΔZ′) may increase a signal-to-noise ratio (SNR) e.g., defined as:

ΔSNR=|ΔI|/I _(n)  (14)

with I_(n) referring to the noise component in the current I. In anotheraspect, increasing the fractional change may reduce dynamic rangerequirements of the current measurement circuit 550.

As non-limiting examples, the fractional change may be increased byoptimizing the design of the sense coil 502 with respect to its geometryand its integration into the wireless power transfer structure (e.g.,wireless power transfer structure 200 with reference to FIGS. 2 and 3),by resonance tuning e.g., using the parallel capacitor 544, and byimproving the Q-factor of the sense circuit 541. Improving the Q-factormay increase the SNR, if the noise current I_(n) is predominantlycircuit intrinsic noise as discussed below with reference to FIG. 5G.

As previously discussed with reference to the circuit 500 of FIG. 5A, itmay be desirable to discriminate between certain categories of objects(e.g., object 110 and 112) e.g., based on the reflected admittanceΔY_(r) that may be indicative of electrical properties of the object110, 112, 114, or vehicle 330.

In some implementations and configurations of the circuit 540 of FIG.5C, the change ΔY in the admittance Y₁₁ caused by an object (e.g.,object 110) is indicative of the reflected admittance ΔY_(r). Therefore,in an aspect of object discrimination, the circuit 540 may be configuredto determine the angle arg{ΔY} and thus the angle arg{ΔY,} with therequired accuracy. However, in some implementations, measuring theadmittance Y₁₁ including the change ΔY may be subject to errors forvarious reasons as previously discussed with reference to the circuit500 of FIG. 5A.

Susceptance compensation in the sense circuit 541 exhibiting a localextremum (minimum or maximum) in the admittance magnitude function|Y_(11,0)(ω) and hence in the resulting current magnitude |I| at themeasurement port 548 provides a mean to calibrate the currentmeasurement circuit 550 and hence the admittance measurement withrespect to the angle arg{ΔY}.

In a first step of an example calibration procedure applicable to theparallel resonant configuration of the circuit 540 of FIG. 5C, the sensefrequency is adjusted to the local minimum of the current magnitude |I|as measured by the uncalibrated current measurement circuit 550supposing absence of a foreign object. At this frequency, the admittanceY_(11,0) and hence the current I at the measurement port 548 may besubstantially real. Otherwise stated, the angles arg{Y_(11,0)} andarg{I} are substantially zero. In a second step of the examplecalibration procedure, the current measurement circuit 550 is correctedby applying a phase shift such that the imaginary part of the complexcurrent value as determined and output by the current measurementcircuit 550 at this frequency vanishes. Applying the phase shift isequivalent to rotating the admittance plane by an angle arg{I_(uncal)}where I_(uncal) refers to the complex current value as determined by theuncalibrated current measurement circuit 550 (before any correction isapplied). This angle correction may be expressed by the followingcomplex multiplication:

I _(cal) =I _(uncal) exp(−j arg{I _(uncal)})  (15)

where I_(cal) refers to the complex current value as determined by thecalibrated current measurement circuit 550.

Applying the angle correction of Equation (15), an object (e.g., object110) reflecting an admittance ΔY_(r) that is imaginary (reactive) mayresult in a measured current change ΔI_(cal) that is substantiallyimaginary. Nevertheless, a residual error may remain in the anglearg{ΔI_(cal)} due to the impact of the series capacitor 546 and theelectrical losses in the sense circuit 541. The residual angle error ofan example parallel resonant configuration of the circuit 540 and for anexample object 110 is provided in TABLE 2.

In some implementations, the residual error is reduced by configuringthe series capacitor 546 with a capacitance C_(s) whose admittanceY_(Cs) is substantially larger (e.g., 10 times larger) than the parallelresonant conductance of the sense circuit 541. In other implementations,the residual error is reduced by computing the error in the measuredangle arg{ΔY} by estimating parameters of the sense circuit 541 (e.g.,the Q-factor) at the actual sense frequency. In further implementations,the residual error is reduced by measuring the admittance Y_(11,0) attwo or more substantially different frequencies and by determining theelements of an equivalent circuit model of the sense circuit 541 (e.g.,the equivalent circuit model illustrated in FIG. 5G) based on themeasured admittances Y_(11,0) employing a best fit method. In someimplementations, these two or more frequencies include at least thefrequency of the minimum and the maximum of |Y_(11,0)(ω)|.

In an implementation configured for series resonance as defined above,the circuit 540 may be configured to measure the impedance Z₁₁ andcorresponding changes ΔZ of Z₁₁ as caused by the object 110, 112, 114,or vehicle 330. In this case, the impedance change ΔZ may be indicativeof the reflected admittance ΔY_(r) as previously introduced. Asdiscussed above with reference to the parallel resonant configuration,the angle arg{ΔZ} may be subjected to an error and therefore may requirecalibration to reduce an error in the measurement of the angle arg{ΔZ}and thus of the angle arg{ΔY_(r)}.

In an implementation configured for series resonance, the circuit 540may be calibrated analogously to the parallel resonant configurationhowever using the local minimum of the impedance function |Z_(11,0)(ω)|where reactance compensation occurs.

In a first step of an example calibration procedure applicable to theseries resonant configuration of the circuit 540 of FIG. 5C, the sensefrequency is adjusted to the local maximum of the current magnitude |I|as measured by the uncalibrated current measurement circuit 550supposing absence of a foreign object. At this frequency, the impedanceZ_(11,0) and hence the current I at the measurement port 548 may besubstantially real. Otherwise stated, the angles arg{Z_(11,0)} andarg{I} are substantially zero. In a second step of the examplecalibration procedure, the current measurement circuit 550 is correctedby applying a phase shift (impedance plane rotation) as given above byEquation (15).

Applying the angle correction of Equation (15), an object (e.g., object110) reflecting an admittance ΔY_(r) that is imaginary (reactive) mayresult in a measured current change ΔI_(cal) that is substantiallyimaginary. Nevertheless, a residual error may remain in the anglearg{ΔI_(cal)} due to the transformation of ΔY_(r) to ΔZ in the lossysense circuit 541. The residual angle error of an example seriesresonant configuration of the circuit 540 and for an example object 110is provided in TABLE 2.

In an example implementation, the residual error due to thetransformation of ΔY_(r) to ΔZ is reduced by measuring the impedanceZ_(11,0) at two or more substantially different frequencies, supposingabsence of a foreign object, and by determining the elements of anequivalent circuit model (e.g., the equivalent circuit model of FIG. 5G)based on the measured impedances Z_(11,0) employing a best fit method.In some implementations, these two or more frequencies include at leastthe frequency of the minimum and the maximum of |Z_(11,0)(ω)|.

The series and the parallel resonant configuration of the circuit 540 ofFIG. 5C are analyzed below with reference to FIG. 5G with respect tovarious characteristics such as the Q-factor, the fractional change, andvarious definitions of SNR based on an equivalent circuit model.

The circuit 560 of FIG. 5D illustrates a further example implementationbased on measuring a complex transimpedance Z₂₁ of a two-port inductivesense circuit (e.g., sense circuit 561, shown in FIG. 5D as the circuitbetween the left and right dashed lines). The transimpedance Z₂₁ ismeasured by applying, from a current source 512, a sinusoidal currentI_(0,1) at the sense frequency with a defined amplitude and phase to themeasurement port 568 (indicated in FIG. 5D by a terminal and a dashedline) and by measuring, using a voltage measurement circuit 510, thecomplex open-circuit voltage V₂ (amplitude and phase) at the measurementport 569 (indicated in FIG. 5D by a terminal and a dashed line). Thetransimpedance Z₂₁ is then determined by dividing the measured voltageV₂ by the defined (known) current I_(0,1).

The sense circuit 561 of FIG. 5D comprises a double-coil inductive senseelement 562 composed of a first (primary) sense coil 562 a having aninductance L₁ and an equivalent series resistance R₁ and a second(secondary) sense coil 562 b having an inductance L₂ and an equivalentseries resistance R₂. FIG. 5D also indicates a mutual inductance L_(M)and an equivalent mutual resistance R_(M) between the first sense coil562 a and the second sense coil 562 b. The equivalent resistances R₁,R2, and R_(M) include a variety of sense element intrinsic andextraneous electrical losses as previously discussed with reference toFIG. 5A. The sense circuit 561 further comprises a first seriescapacitor 564 having a capacitance C_(s,1) electrically connected inseries to the first sense coil 562 a, a second series capacitor 565having a capacitance C_(s,2) electrically connected in series to thesecond sense coil 562 b. The sense circuit 561 further comprises a firstparallel inductor 566 having an inductance L_(p,1) electricallyconnected to the first capacitor 564 and in parallel to the measurementport 568 and a second parallel inductor 567 having an inductance L_(p,2)electrically connected to the second capacitor 565 and in parallel tothe measurement port 569.

Though not indicated in FIG. 5D for purposes of illustration, the seriescapacitors 564, 565 and the parallel inductors 566, 567 may causeelectrical losses that may be represented by respective equivalentseries resistances.

An inductive coupling factor:

k _(L) =L _(M)(L ₁ L ₂)^(−1/2)  (16)

may be defined for the two-port inductive sense element 562. Further, atwo-port inductive sense element (e.g., inductive sense element 562 ofFIG. 5D) may be modeled by a “T”—equivalent circuit based on inductancesL₁, L₂, L_(M) as illustrated in FIG. 5I. Alternatively, a two-portinductive sense element (e.g., inductive sense element 562) may bemodeled by an equivalent circuit illustrated by FIG. 5J comprising theinductances L₁ and L2 in series to respective current-controlled voltagesources:

V _(ind,1) =jωL _(M) I ₂  (17)

V _(ind,2) =jωL _(M) I ₂  (18)

representing the voltage induced into the first and second sense coil,respectively as indicated in FIG. 5J.

In some implementations, the reactance of C_(s,1) substantiallycompensates for the reactance of L₁ providing a local impedance minimum|Z₁₁| (series resonance) substantially at the nominal sense frequency,while the reactance of C_(s,2) substantially compensates for thereactance of L₂ providing a local impedance minimum |Z₂₂| (seriesresonance) substantially at the nominal sense frequency.

In another implementation, the sense circuit 561 is configured toprovide a local minimum of the admittance magnitude functions |Y₁₁(ω)|and |Y₂₂(ω)| (parallel resonance) substantially at the nominal sensefrequency.

In a further implementation, the sense circuit 561 is configured toprovide a local minimum of the admittance magnitude function |Y₁₁(ω)|(parallel resonance) and a local minimum of the impedance magnitudefunction |Z₂₂(ω)| (series resonance) substantially at the nominal sensefrequency.

In yet another implementation, the sense circuit 561 is configured toprovide a local minimum of the impedance magnitude function |Z₁₁(ω)|(series resonance) and a local minimum of the admittance magnitudefunction |Y₂₂(ω)| (parallel resonance) substantially at the nominalsense frequency.

In implementations configured for primary-side and secondary-side seriesresonance, the reactance of the parallel inductors 566 and 567 issubstantially higher than the impedance magnitudes |Z₁₁| and |Z₂₂|,respectively, of the sense circuit 561 at the nominal sense frequency.

In a further example implementation, at least one of the seriescapacitors 564 and 565 is omitted and the sense circuit 561 is operatedas a non-resonant or partially resonant circuit.

In a further aspect, the first series capacitor 564 in combination withthe first parallel inductor 566 form a 2^(nd) order high pass filter toattenuate a low frequency disturbance component in the voltage V₁.Likewise, the second series capacitor 565 in combination with the secondparallel inductor 567 form a 2^(nd) order high pass filter to attenuatea low frequency disturbance component in the voltage V₂ for purposes aspreviously discussed in connection with FIG. 5A.

With reference to FIG. 1, the sense circuit 561, the sense coils 562 aand 562 b, and the respective capacitors 564, 565 and the respectiveinductors 566, 567 may correspond e.g., to the inductive sense circuit106 a, the inductive sense element 107 a (double sense coil), and therespective associated capacitive elements, respectively.

As with the circuit 500 of FIG. 5A, the circuit 560 of FIG. 5D may alsoinclude parasitic capacitances (not shown in FIG. 5D) such as theself-capacitances (intra winding capacitances C_(iw) and intercoilcapacitance), the ground capacitances C_(gnd), and the capacitancesC_(wpt) between each of the sense coils 562 a and 562 b and the WPT coil(e.g., WPT coil 202 of FIG. 2). These capacitances and the associatedelectric stray fields may cause a certain sensitivity of the circuit 560on substantially non-conductive, dielectric objects (e.g., object 112).

Though not shown herein, other transimpedance measurement techniquessuch as the voltage source current measurement technique or any othercombination may apply (e.g., a current source current measurementtechnique). In some implementations (also not shown herein), at leastone of the impedances Z₁₁ and Z₂₂ of the sense circuit 561 isadditionally measured to the transimpedance Z₂₁ (e.g., using one or moreof the techniques as previously discussed with reference to FIG. 5A). Inthese alternative implementations, presence of an object (e.g., object110) is determined based on a change in at least one of an impedanceZ₁₁, Z₂₂, and Z₂₁.

Moreover, at least one of an impedance transformation and balancing mayapply to at least one of the primary-side and secondary-side of thesense circuit 561 (not shown herein). More specifically, with referenceto the circuit 521 of FIG. 53, a transformer (e.g., transformer 526) maybe used instead of the parallel inductors 566 and 567. Alternatively,with reference to the sense circuit 541 of FIG. 5C, a series capacitorand a parallel capacitor (e.g., capacitors 546 and 544, respectively)may apply at least on the primary side.

With reference to FIG. 1, FIG. 5D also illustrates the objects 110, 112,and 114 proximate to the inductive sense element 562. As previouslydiscussed with reference to FIG. 1, presence of the object 110, 112,114, or vehicle 330 may cause a change in one or more electricalcharacteristics of the sense circuit 561. As non-limiting examples, itmay change the self-inductances L₁ and L₂, the equivalent seriesresistances R₁ and R₂, the mutual inductance L_(M), and the mutualequivalent series resistance R_(M) generally resulting in a change ΔZwith respect to the transimpedance Z_(21,0) as measured in absence of aforeign object. Presence of an object (e.g., object 110) may bedetermined if ΔZ satisfies certain criteria (e.g., the magnitude of ΔZexceeds a detection threshold). A change ΔZ in the measured impedanceZ₂₁ may also be caused by a vehicle (e.g., vehicle 330, not shown inFIG. 5D), which may indicate presence of a vehicle above the inductivesense element 562. Further, an impedance change ΔZ may also be caused bya substantially non-conductive, dielectric object (e.g., object 112 or114) proximate to at least one of the sense coils 562 a and 562 b aspreviously discussed with reference to FIG. 1. In other terms, adielectric object (e.g., object 112 or 114) proximate to at least one ofthe sense coils 562 a and 562 b may change one or more of the parasiticcapacitances as mentioned above.

Using a quasi-ideal current source 512, a change ΔZ in thetransimpedance Z₂₁ (e.g., due to presence of the object 110) manifestsin a change ΔV in the voltage V₂ while the current I_(0,1) remainssubstantially unaffected. Therefore, measuring the complex voltage V₂may be equivalent to measuring the complex transimpedance Z₂₁. In otherwords, the complex voltage V₂ may be indicative of the complextransimpedance Z₂₁ and there may be no requirement for additionallymeasuring the current I_(0,1) thus reducing complexity of themeasurement circuit (e.g., measurement circuit 104 of FIG. 1)

In an aspect, it may be useful to define the normalized transimpedancechange of a two-port sense circuit (e.g., sense circuit 561 of FIG. 5D)as:

ΔZ′=(Z ₂₁ −Z _(21,0))/|Z _(21,0) |=ΔZ/|Z _(21,0)|  (19)

and, correspondingly, the normalized transadmittance change as:

ΔY′=(Y ₂₁ −Y _(21,0))/|Y _(21,0) |=ΔY/|Y _(21,0)|  (20)

also referred herein as to the fractional change. As with the circuit501 of FIG. 5A, the fractional change ΔZ′ (or ΔY′) caused by a definedtest object (e.g., object 110) placed at a defined position relative tothe inductive sense element 562 may relate to the detection sensitivityof an object detection circuit (e.g., the multi-purpose detectioncircuit 100 of FIG. 1) based on a two-port inductive sense circuit(e.g., sense circuit 561). Increasing the fractional change ΔZ′ (or ΔY′)may increase a detection sensitivity of the circuit 560. Morespecifically, it may increase a signal-to-noise ratio (SNR) e.g.,defined as:

ΔSNR _(V) =ΔV|/V _(n)  (21)

with V_(n) referring to the noise component in the voltage V₂.

As non-limiting examples, the fractional change may be increased byoptimizing the design and arrangement of the sense coils 562 a and 562b, their integration into the wireless power transfer structure (e.g.,wireless power transfer structure 200 with reference to FIGS. 2 and 3),by resonance tuning e.g., using the series capacitors 564 and 565 aspreviously described, and by improving a Q-factor of the sense circuit561.

In an example implementation, the fractional change ΔZ′ (or ΔY′) issubstantially increased by configuring and arranging the sense coils 562a and 562 b such that the mutual inductance L_(M) substantially vanishesin absence of a foreign object, resulting in a transimpedance |Z_(21,0)|that is substantially zero. Example implementations of double sense coilarrangements providing a substantially zero mutual inductance L_(M) aredescribed in U.S. patent application Ser. No. 16/358,534, titled ForeignObject Detection Circuit Using Mutual Impedance Sensing, the entirecontents of which are hereby incorporated by reference.

The circuit 580 of FIG. 5E illustrates yet a further exampleimplementation based on measuring a complex transimpedance Z₂₁ of atwo-port inductive sense circuit 581 (shown in FIG. 5E as the circuitbetween the left and the right dashed line). The transimpedance Z₂₁ ismeasured by applying, from a current source 512, a sinusoidal currentI_(0,1) at the sense frequency with a defined amplitude and phase to themeasurement port 588 (indicated in FIG. 5E by a terminal and a dashedline) and by measuring, using a voltage measurement circuit 510, thecomplex open-circuit voltage V₂ (amplitude and phase) at the measurementport 589 (indicated in FIG. 5E by a terminal and a dashed line). Thetransimpedance Z₂₁ is then determined by dividing the measured voltageV₂ by the defined (known) current I_(0,1).

The sense circuit 581 of FIG. 5E comprises a double-coil inductive senseelement 562 with reference to FIG. 5D composed of the first sense coil562 a having an inductance L₁ and an equivalent series resistance R₁ anda second sense coil 562 b having an inductance L₂ and an equivalentseries resistance R₂. FIG. 5E also indicates the mutual inductance L_(M)and the equivalent mutual resistance R_(M). The equivalent resistancesR₁, R₂, and R_(M) include a variety of sense element intrinsic andextraneous electrical losses as previously discussed with reference toFIG. 5A. The sense circuit 561 further comprises a series capacitor 584having a capacitance C_(s) electrically connected to the second terminalof the sense coils 562 a and 562 b, a first parallel inductor 586 havingan inductance L_(p,1) electrically connected to the first terminal ofthe sense coil 562 a and in parallel to the measurement port 588 and asecond parallel inductor 587 having an inductance L_(p,2) electricallyconnected to the first terminal of the sense coil 562 b and in parallelto the measurement port 589. The circuit 580 further illustrates thesense signal current source 512 and the voltage measurement circuit 510electrically connected to the measurement ports 588 and 589,respectively.

Though not indicated in FIG. 5E for purposes of illustration, the seriescapacitor 584 and the parallel inductors 586, 587 may cause electricallosses that may be represented by respective equivalent seriesresistances.

In an example implementation, the sense coils 562 a and 562 b aretightly coupled resulting in an inductive coupling factor k_(L) asdefined by Equation (16) that is near unity (k_(L)≈<1). Exampleimplementations of double-coil inductive sense elements 562 providing aninductive coupling factor k_(L) near unity are described in U.S. patentapplication Ser. No. 16/358,534, titled Foreign Object Detection CircuitUsing Mutual Impedance Sensing, the entire contents of which are herebyincorporated by reference.

The sense circuit 581 may be configured to provide a local minimum inthe transimpedance magnitude function |Z_(21,0)(ω)| (series resonance)substantially at a nominal sense frequency. Alternatively, the sensecircuit 581 may be configured to provide a local minimum in thetransadmittance magnitude function |Y_(11,0)(ω)| substantially at thenominal sense frequency.

In an example series resonant configuration of the sense circuit 581using an inductive sense element 562 with k_(L)≈<1, the reactance of theseries capacitor 584 substantially compensates for the reactance of themutual inductance L_(M) providing a local minimum in the transimpedancemagnitude function |Z_(21,0)(ω)| (series resonance) substantially at thenominal sense frequency. The principle of mutual reactance compensationmay become more evident by contemplating FIG. 5I illustrating a “T”equivalent circuit model 562-1 of the two-port inductive sense element562 and by considering the capacitance C_(s) of the capacitor 584inserted in series to the mutual inductance L_(M). With k_(L)≈<1, boththe series inductances L₁-L_(M) and L₂-L_(M) become substantially zero.

In this series resonant configuration, the inductance L_(p,1) andL_(p,2) of the parallel inductor 586 and 587, respectively, may besimilar or larger than the inductance L₁ and L₂ of the sense coils 562 aand 562 b, respectively. Stated in other terms, the impedance magnitudeof the parallel inductor 586 and 587 may be substantially higher thanthe impedance magnitude |Z₁₁| and |Z₂₂|, respectively, of the sensecircuit 581 at the nominal sense frequency. In this configuration, theparallel inductors 586 and 587 may exert a negligible impact on theimpedances and transimpedance |Z₁₁|, |Z₂₂|, and |Z₂₁|, respectively, atthe nominal sense frequency.

In an example parallel resonant configuration of the sense circuit 581using an inductive sense element 562 with k_(L)≈<1, the reactance of theseries capacitor 584 overcompensates for the reactance of the mutualinductance L_(M) at the nominal sense frequency. The residual capacitivesusceptance of the series connection of the capacitor 584 and the mutualinductance L_(M) is substantially compensated for by the susceptance ofthe parallel inductors 586 and 587 providing a transadmittance Y_(21,0)that is substantially real (resistive). In this configuration, theinductances L_(p,1) and L_(p,2) of the parallel inductors 586 and 587,respectively, may be smaller, similar, or larger than the inductance L₁and L₂ of the sense coils 562 a and 562 b, respectively. Stated in otherterms, the admittance magnitude of each of the parallel inductors 586and 587 may be substantially (e.g., 20 times) higher than the admittancemagnitudes |Y₁₁| and |Y₂₂|, respectively, as presented at the nominalsense frequency. In this configuration, the parallel inductors 586 and587 exert a significant impact on the admittance and transadmittancemagnitudes |Y₁₁|, |Y₂₂|, and |Y₂₁|, respectively, at the nominal sensefrequency.

In some implementations, the parallel inductors 586 and 587 togetherwith the series capacitor 584 are used for purposes of resonance tuningand transimpedance transformation, e.g., to transform the transimpedanceZ₂₁ to match the sense circuit 581 with an operating transimpedancerange as previously mentioned with reference to FIG. 1. The inductanceratios L₁/L_(p,14) and L₂/L_(p,2) may be parameters to control theimpedance magnitudes |Z_(11,0)|, Z_(22,0)|, and Z_(21,0)|.

Impedance and transimpedance transformation may be particularlyeffective, if the sense circuit 581 is configured for parallelresonance. More specifically, increasing the inductance ratiosL₁/L_(p,1) and L₂/L_(p,2), while maintaining parallel resonance at thenominal sense frequency, may substantially increase the admittancemagnitudes |Y_(11,0)|, Y_(22,0)|, and |Y_(21,0)| of the parallelresonant configuration at the nominal sense frequency.

increasing the inductance ratios L₁/L_(p,1) and L₂/L_(p,2), whilemaintaining resonance at the nominal sense frequency, may also somewhatdecrease the impedance magnitudes |Z_(11,0)|, |Z_(22,0)|, and |Z_(21,0)|as presented at the nominal sense frequency in the series resonantconfiguration of the sense circuit 581. However, impedancetransformation may be limited and far less effective than that of theparallel resonant configuration.

In a further aspect, the series capacitor 584 in combination with thefirst parallel inductor 586 form a 2^(nd) order high pass filter toattenuate a low frequency disturbance component in the voltage V₁.Likewise, the second series capacitor 584 in combination with the secondparallel inductor 587 form a 2^(nd) order high pass filter to attenuatea low frequency disturbance component in the voltage V₂ for purposes aspreviously discussed in connection with FIG. 5A.

With reference to FIG. 1, the sense circuit 581, the sense coils 562 aand 562 b, and the capacitor 584 may correspond e.g., to the inductivesense circuit 106 a, the inductive sense element 107 a (double sensecoil), and the respective associated capacitive element, respectively.

With reference to FIG. 1, FIG. 5E also illustrates the objects 110, 112,and 114 proximate to the inductive sense element 562. As previouslydiscussed with reference to FIG. 1, presence of the object 110, 112,114, or vehicle 330 may cause a change in one or more electricalcharacteristics of the sense circuit 561 as previously discussed withreference to FIG. 5D. As non-limiting examples, it may change theself-inductances L₁ and L₂, the equivalent series resistances R₁ and R₂,the mutual inductance L_(M), and the equivalent mutual resistance R_(M)generally resulting in a change ΔZ with respect to the transimpedanceZ_(21,0) as measured in absence of a foreign object. In animplementation with k_(L)≈<1, the change ΔZ may be primarily related toa change in the mutual inductance L_(M) and the equivalent mutualresistance R_(M).

As with the sense circuit 561 of FIG. 5D, the fractional change ΔZ′ (orΔY′) caused by a defined test object (e.g., object 110) may relate tothe detection sensitivity of the sense circuit 581. It may beappreciated that using an inductive sense element 562 with k_(L)≈<1reduces the impact of the equivalent series resistances R₁ and R₂ on thefractional change, if compared e.g., to the circuit 500 of FIG. 5A. Thefractional change of the sense circuit 581 with k_(L)≈<1 is governed bythe Q-factor:

Q _(M) ≈ωL _(M) /R _(M)  (22)

of the inductive sense element 562 with respect to L_(M) and R_(M).

Moreover, the impedance change ΔZ may reflect electrical properties ofthe object 110, 112, or 114 as discussed with reference to the circuit500 of FIG. 5A.

The circuit 580 of FIG. 5E may further allow calibration to reduce anerror in the measurement of the angle arg{ΔZ} by applying a procedure aspreviously described with reference to FIG. 5A.

FIGS. 5F and 5G illustrate an equivalent circuit model 500-1 and 540-1,respectively, used below for purposes of a theoretical analysis andperformance comparison. More specifically, the equivalent circuit model500-1 is used to analyze the circuit 500 of FIG. 5A and the circuit 520of FIG. 5B (using the transformer 526), while the equivalent circuitmodel 540-1 serves for the analysis of the circuit 540 of FIG. 5C. Eachof the circuits 500, 520, and 540 are analyzed with respect to itsseries and parallel resonant configuration and with respect to variouscharacteristics such as the impedance and the Q-factor of the sensecircuit at resonance, the fractional change, and various SNRs as definedbelow.

For purposes of comparison, an identical sense coil 502 an equal sensecoil current level |I_(L)| is assumed for both configurations of thecircuits 500, 520, and 540, though practical implementations configuredfor parallel resonance may prefer a sense coil 502 with a lowerinductance L. Comparing SNRs at the same sense coil current level|I_(L)| may be meaningful e.g., if the current level |I_(L)| is emissionor power constraint. Further, it is assumed that the circuits in bothconfigurations are adjusted to a common resonant frequency substantiallycorresponding with the nominal sense frequency that is substantiallyhigher than the WPT operating frequency.

The equivalent circuit model 500-1 as illustrated in FIG. 5F comprisesthe sense coil's 502 inductance L and its equivalent series resistanceR, the series capacitor's 504 capacitance C_(s) the parallel inductor's506 inductance L_(p) and its equivalent series resistance R_(Lp), anideal sense signal current source 512 and an ideal voltage measurementcircuit 510. It may be appreciated that in practical implementations,losses in capacitors are generally substantially lower than losses ininductors. Therefore, the equivalent series resistance of seriescapacitor 504 is neglected (not shown) in the equivalent circuit model500-1 of FIG. 5F. Further, the equivalent circuit model 500-1 includesan impedance ΔZ_(r) in series to the inductance L representing thereflected impedance of the object 110, 112, or 114 proximate to thesense coil 502. (The reflected impedance ΔZ_(r) may be regarded as theobject 110, 112, or 114 as illustrated in FIG. 5A abstracted away). Theequivalent circuit model 500-1 also includes a noise voltage sourceV_(sn) in series to the inductance L representing the noise voltageinductively and capacitively coupled into sense coil 502 by the magneticand electric field as generated when WPT is active. The noise voltageV_(sn) may include any low frequency component (e.g., at the fundamentalof the WPT operating frequency and harmonics thereof) as well as anyhigh frequency component (e.g., switching noise at the sense frequency).The equivalent circuit model 500-1 further indicates the impedance Z₁₁and the admittance Y₁₁ (=1/Z₁₁), the drive current I₀ with an additivenoise current component I_(0,n), the sense signal voltage V with anadditive noise voltage V_(n), and the measurement port 508 (indicated bythe terminal and the dashed line) where the current I₀+I_(o,n) isapplied, the voltage V+V_(n) is measured, and where Z₁₁ or Y₁₁ refer to.Because the equivalent circuit model 500-1 applies to the circuit 500 ofFIG. 5A or the circuit 520 of FIG. 5B, the reference numerals 500 and520, respectively, are used instead in the following theoreticalanalysis.

To analyze the series and parallel resonant configuration of the circuit500 of FIG. 5F, the following assumptions:

ωL >>R  (23)

ωL_(p)>>R_(Lp)  (24)

|ΔZ |<<R  (25)

are made for a frequency range about the resonant frequency.

In an implementation configured for series resonance and with areactance:

ωL _(p) >>|Z ₁₁|  (26)

in a frequency range about the series resonant frequency, the impedanceZ₁₁ at the measurement port 508 of the circuit 500 of FIG. 5F inpresence of an object (e.g., object 110) may be expressed as:

Z ₁₁ ≈R+(jωC_(s))⁻¹ +jωL+ΔZ _(r)  (27)

In absence of a foreign object, a local minimum of |Z_(11,0)(ω)| (seriesresonance) occurs substantially at an angular frequency ω satisfying:

(jωC _(s))⁻¹ +jωL≈0  (28)

yielding the series resonant angular frequency:

ω_(s)≈(LC _(s))^(−1/2)  (29)

At this frequency, the impedance Z_(11,0) becomes substantially real:

Z _(11,0) ≈Re{Z _(11,0) }=R _(s) ≈R  (30)

with R_(s) denoting the series resonant resistance, while the impedanceZ₁₁ in presence of an object (e.g., object 110) is approximately:

Z₁₁ ≈R _(s) +ΔZ≈R+ΔZ _(r)  (31)

with ΔZ_(r) referring to the reflected impedance as previously definedwith reference to FIG. 5A.

Applying Equations (30) and (31) to Equation (8), the fractional changeΔZ′ for the series resonant configuration of the circuit 500 of FIG. 5Fbecomes approximately:

ΔZ′≈ΔZ _(r) /R _(s) ≈ΔZ _(r) /R  (32)

Using the definition of Equation (6) of the normalized reflectedimpedance ΔZ_(r)′ at ω_(s) and defining a Q-factor of the seriesresonant configuration of the circuit 500 of FIG. 5F:

Q_(s)≈ω_(s)L/R_(s)  (33)

which approximately equals the Q-factor of the sense coil 502 at theseries resonant frequency:

Q=ω _(s) L/R≈Q _(s)  (34)

the fractional change may also be written in terms of ΔZ_(r)′ and Q_(s)as:

ΔZ′≈Q_(s)ΔZ_(r)′  (35)

Equation (35) shows that reactance compensation in the series resonantconfiguration of the circuit 500 of FIG. 5F multiplies the normalizedreflected impedance ΔZ_(r)′ by the Q-factor Q_(s) that approximatelyequals the Q-factor Q of the sense coil 502.

To analyze the parallel resonant configuration of the circuit 500 ofFIG. 5F, the additional assumption:

|ωL−(ωC _(s))⁻¹ |>>R  (36)

is made for a frequency range about the resonant frequency. Theadmittance Y₁₁ at the measurement port 508 in presence of an object(e.g., object 110) may be expressed as:

Y ₁₁=(R _(Lp) +jωL _(p))⁻¹+(R+jωL+(jωC _(s))⁻¹ +ΔZ _(r))⁻¹  (37)

Using Equations (23), (24), (25), (36) and the approximation:

1/(1+x)≈1−x  (38)

valid for |x|<<1, where x may be a complex number, and neglectinginsignificant terms, the admittance Y₁₁ of Equation (37) may beapproximated as:

Y ₁₁=(jωL _(p))⁻¹ +R _(Lp)(ωL _(p))⁻²+(jωL+(jωC _(s))⁻¹)⁻¹+(R+ΔZ _(r))(ωL−(ωC _(s))⁻¹)⁻²  (39)

In absence of a foreign object, a local minimum of |Y_(11,0)(ω)|(parallel resonance) occurs substantially at an angular frequency ωsatisfying:

(jωC _(s))⁻¹ +jω(L+L _(p))≈0  (40)

yielding for the parallel resonant angular frequency:

ω_(p)≈(C _(s)(L+L _(p)))^(−1/2)  (41)

At this frequency, the admittance Y_(11,0) becomes substantially real:

Y _(11,0) ≈Re{Y _(11,0) }=G _(p)=(R+R _(Lp))/(ω_(p) L _(p))²  (42)

with G_(p) denoting the parallel resonant conductance, while theadmittance Y₁₁ in presence of an object (e.g., object 110) isapproximately:

Y ₁₁ ≈G _(p) +ΔY≈(R+R _(Lp) +ΔZ _(r))/(ω_(p) L _(p))²  (43)

where:

ΔY≈ΔZ _(r)/(ω_(p) L _(p))²  (44)

defines the admittance change due to the object.

Further, defining the Q-factor of the sense coil 502:

Q=ω _(p) L/R  (45)

and the Q-factor of the parallel inductor 506:

i Q_(Lp)=ω_(p) L _(p) /R _(Lp)  (46)

the inductance ratio:

n _(L) =L/L _(p)  (47)

the admittance Y_(11,0) of Equation (42) at ω_(p) may be expressed as:

Y _(11,0) ≈G _(p) ≈n _(L)(Q/Q _(Lp))+n _(l))/(Qω _(p) L)  (48)

For the case Q_(Lp)=Q and n_(L)>>1, the parallel resonant conductanceG_(p) becomes approximately:

G _(p) ≈n _(l) ²/(Qω _(p) L)  (49)

According to Equation (48), the admittance Y₁₁ at ω_(p) of the sensecircuit 501 of FIG. 5F can be modified (e.g., decreased) by adjustingthe inductance ratio n_(L)=L/L_(p) accordingly, while maintainingparallel resonance substantially at the nominal sense frequency.Therefore, in some implementations, the parallel resonant configurationof the circuit 500 of FIG. 5A is employed as an alternative to using atransformer (e.g., transformer 526 of FIG. 5B) for transforming theadmittance Y₁₁ to be within a suitable operating range as previouslydiscussed with reference to FIG. 5B.

Applying Equations (42) and (44) to Equation (9), the fractional changeΔY′ for the parallel resonant configuration of the circuit 500 of FIG.5F becomes approximately:

ΔY′=ΔY/G _(p) ≈ΔZ _(r)/(R+R _(Lp))  (50)

showing that the admittance change ΔY is substantially proportional tothe reflected impedance ΔZ_(r). Therefore, the angle arg{ΔY} of themeasured admittance change ΔY is indicative of the angle arg{ΔZ_(r)}. Aspreviously described with reference the circuit 500 of FIG. 5A, theaccuracy of the measured angle may be improved by applying calibration.

Defining the Q-factor of the parallel resonant configuration of thesense circuit 501 of FIG. 5F:

Q _(p)=ω_(p)(L _(p) +L)/(R+R _(Lp))≈n _(L)(1+n _(L))/(G _(p)ω_(p)L)  (51)

which may be also expressed in terms of the Q-factors Q and Q_(Lp) as:

Q _(p) =Q(1+n _(L)_/((Q/Q _(Lp))+n _(l))  (52)

using the definition of Equation (6) of the normalized reflectedimpedance at ω_(p), and applying Equations (51) and (47) to Equation(50), the fractional admittance change ΔY′ may also be written as:

ΔY′≈Q _(p) ΔZ _(r) ′n _(L)/(1+n _(L))  (53)

For the case Q_(Lp)=Q, the fractional change becomes:

ΔY′=QΔZ _(r) ′n _(L)/(1+n _(L))  (54)

and for Q_(Lp)>>Q:

ΔY′≈ZΔZ_(r)′  (55)

The fractional change |ΔY′| of the parallel resonant configuration ofthe circuit 500 of FIG. 5F as given by Equation (53) is generally lowerthan |ΔZ′| of the series resonant configuration as given by Equation(35) but approaches |ΔZ′| as the inductance ratio n_(L) or the Q-factorQ_(Lp) increases. In an example implementation configured with L_(p)=L(n_(L)=1) and Q_(Lp)=Q, the fractional change |ΔY′| of the parallelresonant configuration is approximately ½ of the fractional change |ΔZ′|of the series resonant configuration, while in another exampleimplementation with L_(p)=L/4(n_(L)=4) and Q_(Lp)=Q, the fractionalchange |ΔY′| amounts to about ⅘ of |ΔZ′|.

In a further aspect, the drive current level I₀, the resulting voltage Vat the measurement port 508, and the drive power level P are considered.In some implementations based on the circuit 500 of FIG. 5F, the currentlevel I₀ of the current source 512 is adjusted to achieve a specifiedcurrent level |I_(L)| in the sense coil 502. For the series resonantconfiguration of the circuit 500 of FIG. 5F, the current level I₀approximately equals |I_(L)|:

I ₀ ≈|I _(L)|  (56)

resulting in a voltage across the measurement port 508:

V≈|Z _(11,0) |I ₀ ≈R|I _(L|)  (57)

and in a drive power level:

P≈VI ₀ ≈R|I _(L|) ²  (58)

Using Equations (42), (47), and (51) for the parallel resonantconfiguration of the circuit 500 of FIG. 5F, it can be shown that thecurrent |I_(L)| through the sense coil 502 at parallel resonance isapproximately Q_(p)/(1+n_(L)) times higher than the drive current levelI₀ providing:

I ₀ ≈|I _(L)|(1+n _(L))/Q _(p) ≈|I _(L)|((Q/Q _(Lp))+n _(L))/Q  (59)

The voltage across the measurement port 508 becomes approximately:

V≈I ₀ /|Y _(11,0) |≈I ₀ /Q _(p) ≈|I _(L)|/(QGn _(L))  (60)

and the drive power:

P≈I ₀ V≈(|I _(L)|² /G) ((Q/Q _(Lp))+n _(L))/(Q ² n _(L))  (61)

In a further aspect, the SNR in the voltage Vat the measurement port 508may be considered. As with the fractional change, the SNR may determinethe sensitivity of the multi-purpose detection circuit 100. It may bedistinguished between the intrinsic SNR (the sense signal-to-circuitintrinsic noise ratio) and the extrinsic SNR (the sensesignal-to-circuit extrinsic noise ratio). With reference to FIG. 5F,circuit intrinsic noise may include contributions from the noise currentI_(0,n) caused by the current source 512 and from noise inherent to thevoltage measurement circuit 510. Further, it may include a contributionfrom thermal noise of the loss resistances R and R_(Lp) inherent to thesense circuit 501. Circuit extrinsic noise may include any disturbancesignal component inductively and capacitively coupled into the sensecoil 502 (e.g., via the magnetic and electric field as caused by the WPTsystem when active). In some implementations, circuit extrinsic noisemay prevail when WPT is active, while circuit intrinsic noise maydetermine the SNR when WPT is inactive. As previously mentioned, incertain implementations and use cases, the multi-purpose detectioncircuit 100 is also used when WPT is inactive (e.g., for determiningpresence of a foreign object, a vehicle, a type of vehicle, or theposition of a vehicle).

It may be further distinguished between a narrowband SNR resulting atthe nominal sense frequency in the bandwidth of the voltage measurementcircuit 510 and a broadband SNR defined in a larger bandwidth e.g., alsocovering the WPT operating frequency. The former mainly relates to thesensitivity of a multi-purpose detection circuit 100, while the lattermay determine the dynamic range and filtering requirements of thevoltage measurement circuit 510.

In another aspect, it may be meaningful to define the narrowband SNR atthe measurement port 508 of the circuit 500 of FIG. 5F as given byEquation (10), where |ΔV| denotes the magnitude of the change in themeasured voltage V due to the presence of an object (e.g., object 110)and V_(n) the additive noise voltage component as indicated in thecircuit 500 of FIG. 5F. More specifically, the voltage change |ΔV| mayrefer to the r.m.s. voltage and V_(n) to the r.m.s. noise voltage asmeasured at the nominal sense frequency in the bandwidth B_(m) of thevoltage measurement circuit 510. This noise voltage V_(n) may includecircuit intrinsic and extrinsic noise components as discussed above. TheSNR as given by Equation (10) is referred herein as to the differentialnarrowband SNR.

In yet a further aspect, it may be meaningful to define the broadbandextrinsic SNR at the measurement port 508 of the circuit 500 of FIG. 5Fas:

SNR _(w) =|V|/V _(W)  (62)

where |V| denotes the magnitude of the sense signal voltage and V_(W)the disturbance voltage at the fundamental WPT operating frequency,which may be a prominent component in V_(n) when WPT is active. Morespecifically, the voltage |V| may refer to the r.m.s. voltage of thesense signal and V_(W) to the r.m.s. disturbance voltage as measured atthe measurement port 508 at the fundamental WPT operating frequency.

Using Equation (19), the differential narrowband extrinsic SNR for theseries resonant configuration of the circuit 500 of FIG. 5F may beexpressed as:

ΔSNR _(ex,s) ≈|ΔZ _(r) ||I _(L) /V _(sn) ≈|ΔZ _(r) ′||V _(L) |/V _(sn)=|ΔZ _(r)′|ω_(s) L|I _(L) |/V _(sn)  (63)

with |I_(L)| denoting the magnitude of the sense signal current in thesense coil 502, which approximately equals the source current level I₀,and V_(sn) the noise voltage as indicated in FIG. 5F.

Since the sense circuit 501 transforms the voltage drop across ΔZ_(r) toΔV in the same way as it transforms V_(sn) to V_(n), Equation (21) mayalso apply to the parallel resonant configuration of the circuit 500,meaning that:

ΔSNR _(ex,p)=(|I_(L) |/V _(sn))ω_(s) L|ΔZ _(r)′|  (64)

Equation (63) and (64) show that the differential narrowband extrinsicSNR for the circuit 500 of FIG. 5F is no function of the Q-factor forboth the series and the parallel resonant configuration.

In some implementations (e.g., where the sense signal is numericallygenerated and converted to an analog signal using a digital-to-analogconverter (DAC) e.g., in the signal generator circuit 406 with referenceto FIG. 4), the noise current I_(0,n) as indicated in FIG. 5F may causethe predominant contribution in V_(n) when WPT is inactive. In thiscase, the noise voltage V_(n) for the series resonant configuration isapproximately:

V_(n)‥RI_(0,n)  (65)

while the voltage change |ΔV| in presence of an object (e.g., object110) is:

|ΔV|≈|I _(L) ||ΔZ _(r) |≈|I ₀ ||ΔZ _(r)|  (66)

Applying Equations (35), (65), and (66) to Equation (10), thedifferential narrowband intrinsic SNR with respect to the noise currentI_(0,n) for the series resonant configuration of the circuit 500 of FIG.5F may be expressed as:

ΔDNR _(int,s)≈(|I ₀ |/I _(0,n))|ΔZ _(r) |/R  (67)

Using Equation (33), Equation (67) may also be written in terms of theQ-factor Q_(s) and the normalized reflected impedance ΔZ_(r)′ as:

ΔSNR _(int,s)≈(|I ₀ |/I _(0,n))Q _(s) |ΔZ _(r)′|  (68)

Using:

ΔY′<<1  (69)

which follows from Equations (23) and (25) and Equation (38), themagnitude |ΔV| of the voltage change in the parallel resonantconfiguration of the circuit 500 of FIG. 5F may be approximated as:

|ΔV|=|(I ₀ /Y ₁₁)−(I ₀ /Y _(11,0))|=|I ₀||(Y _(11,0) +ΔY)^('1) −Y_(11,0))⁻¹ 51 ≈|I ₀ ||ΔY|/|Y _(11,0)|²  (70)

With the noise current I_(0,n) as the predominant contribution, thenoise voltage V_(n) at the parallel resonant frequency becomes:

V _(n) =I _(0,n) /|Y _(11,0)|  (71)

Applying Equations (70), (71), and (50) to Equation (10), thedifferential narrowband intrinsic SNR with respect to the noise currentI_(0,n) for the parallel resonant configuration of the circuit 500 ofFIG. 5F may be expressed as:

ΔSNR _(int,p)≈(|I ₀ |/I _(0,n))|ΔY′|≈(|I ₀ |/I _(0,n))|ΔZ _(r)|/(R _(Lp)+R)  (72)

Using Equation (53), Equation (72) may also be written in terms of theQ-factor Q_(p) and the normalized reflected impedance ΔZ_(r)′ as:

ΔSNR _(int,p)≈(|I ₀ |/I _(0,n))|ΔZ _(r) ′|Q _(p) n _(L)/(1+n _(L))  (73)

Similar considerations may be made for the thermal noise though likelyless significant in practical implementations as further shown belowwith reference to TABLE 2. As previously mentioned, a thermal noisevoltage is generated by the series equivalent loss resistances R_(Lp)and R. The noise voltage component V_(n) at the series resonantfrequency may be considered as the thermal noise voltage generated bythe series resonant resistance R_(s) as defined by Equation (30) andbecomes approximately:

V _(n)=(4kTB _(m) R _(s))^(1/2)≈(4kTB _(m) R)^(1/2)  (74)

where k denotes the Boltzmann constant, T the absolute temperature ofthe sense coil 502, and B_(m) the equivalent noise bandwidth of thevoltage measurement circuit 510. Applying Equation (66) and (74) toEquation (10) provides for the differential narrowband intrinsic SNRwith respect to thermal noise for the series resonant configuration ofthe circuit 500 of FIG. 5F:

ΔSNR _(int,s) ≈|I _(L) ||ΔZ _(r) |/V _(n) ≈|I _(L)|ω_(s) L|ΔZ_(r)′|/(4kTB _(m) R)^(1/2)  (75)

Accordingly, the thermal noise voltage V_(n) as resulting at parallelresonance may be considered as the thermal noise generated by theparallel resonant conductance G_(p) as defined by Equation (42).Assuming equal temperature T for the sense coil 502 and the parallelinductor 506, the noise voltage V_(n) becomes approximately:

V _(n≈()4kTB _(m) /G _(p))^(1/2)  (76)

Using Equation (70), (50), (42), (51), and the relation:

|I ₀ |=|V|G _(p) ≈I _(L)|ω_(p) L _(p) G _(p)  (77)

the voltage change ΔV may be expressed as:

|ΔV|≈|I ₀ ||ΔY|/G _(p) ² ≈|I _(L)|ω_(p) L _(p) |ΔY′|≈|I _(L)|ω_(p) L_(p) |ΔZ _(r)|/(R+R _(Lp))  (78)

Applying Equations (76) and (78) to Equation (10) also using Equation(42), provides for the differential narrowband intrinsic SNR withrespect to the thermal noise for the parallel resonant configuration ofthe circuit 500 of FIG. 5F:

ΔSNR _(int,p) ≈|I _(L)|ω_(p) L|ΔZ _(r)′|/(4kTB _(m)(R+R_(Lp)))^(1/2)  (79)

In a further aspect, the broadband extrinsic SNR as defined by Equation(62) with respect to the induced voltage component V_(sW) at thefundamental WPT operating angular frequency ω_(W) is considered.Assuming the magnetic field coupling as the predominant contribution,the disturbance signal voltage V_(sn) may relate to the WPT coil currentI_(WPT) as follows:

V_(sn)≈V_(sW)≈ω_(W)L_(sW)I_(WPT)  (80)

where L_(sW) denotes the mutual inductance between the sense coil 502and the WPT coil (e.g., WPT coil 202 with reference to FIGS. 2 and 3).Further, assuming:

1/(ω_(W) C _(s))>>ω_(W) L  (81)

ω_(s)>>ω_(W)  (82)

and using Equation (29) and (47), the disturbance voltage componentV_(W) in the voltage V for the series resonant configuration of thecircuit 500 of FIG. 5F becomes approximately:

V _(n) =V _(W) ≈V _(W)ω_(W) C _(s)ω_(W) L _(p) ≈V _(sW)(ω_(W)/ω_(s))² /n_(L)  (83)

The factor (ω_(W)/ω_(s))²/n_(L) may be considered as the attenuation ofthe low frequency induced voltage V_(sW) by the high pass filter effectof the sense circuit 501. Using:

|V|=|I ₀ |R _(s) ≈|I _(L) |R  (84)

and applying Equations (33), (83), (80), (84), and (47) to Equation(62), the broadband extrinsic SNR for the series resonant configurationof the circuit 500 of FIG. 5F may be expressed in terms of the Q-factorQ_(s) and the inductance ratio n_(L) as:

SNR _(W,s)≈(|I _(L) |/V _(sW))ω_(s) L(ω_(s)/ω_(W))² n _(L) /Q _(s)  (85)

Using Equation (41), (81), and (82), the disturbance voltage componentV_(W) in the voltage V for the parallel resonant configuration of thecircuit 500 of FIG. 5F becomes approximately:

V _(n) =V _(W) ≈V _(sW)ω_(W) C _(s)ω_(W) L _(p) ≈V_(sW)(ω_(W)/ω_(p))²/(1+n _(L))  (86)

The factor (ω_(W)/ω_(p))²/(1+n_(L)) may be considered as the attenuationof the low frequency induced voltage V_(sW) by the high pass filtereffect of the sense circuit 501. Further, expressing the sense signalvoltage |V| at the angular frequency co_(p) in terms of the sense coilcurrent |I_(L)| using Equation (40):

|V|≈|I _(L)|((ω_(p) C _(s))⁻¹−ω_(p) L)≈|I _(L)|ω_(p) L _(p)  (87)

and applying Equations (86) and (87) to Equation (62), the broadbandextrinsic SNR with respect to the WPT fundamental disturbance voltagecomponent V_(sW) for the parallel resonant configuration of the circuit500 of FIG. 5F may be expressed as:

SNR _(W,p)≈(|I _(L) |/V _(sW))ω_(p) L(ω_(p) /ω _(W))²(1+n _(L))/n_(L)  (88)

In yet another aspect, the temperature sensitivity as defined byEquations (11) and (12) for the real and imaginary part of Z₁₁,respectively, is considered. Using Equation (35), the real parttemperature sensitivity of the circuit 500 of FIG. 5F may be expressedas:

S_(ϑ, R) =Re{αZ′ _(ϑ) }/Re{ΔZ′}≈Re{ΔZ′ _(ϑ)}/(Q _(s) Re{ΔZ _(r)′})  (89)

Equation (89) shows that the real part temperature sensitivity reducesas the Q-factor Q_(s) of the sense circuit 501 increases. However, theimaginary part temperature sensitivity may not improve and may onlyreduce by lowering a temperature coefficient associated with theinductive and capacitive elements of the sense circuit 501.

In some implementations, components and materials with a low temperaturecoefficient (e.g., NP0-type capacitors) are used. In otherimplementations, temperature sensitivity is reduced e.g., using acombination of components or materials with a positive temperaturecoefficient and components or materials with a negative temperaturecoefficient in a manner such that the overall thermal drift is cancelledout.

Equations (8) to (89) may also apply to the circuit 520 of FIG. 5B withsome minor modifications e.g., by replacing the inductance L by L+L_(σ),the series resistance R by R+R_(w), the inductance L_(p) by L_(m), andthe series resistance R_(Lp) by R_(m), where L_(σ) denotes thetransformer's 526 secondary referred leakage inductance, R_(w) itssecondary referred equivalent series resistance with respect to theconductor losses, L_(m) its secondary referred main inductance, andR_(m) its secondary referred equivalent series resistance with respectto the core losses with reference to FIG. 5H. Further, if L_(σ) is asubstantial portion of L+L_(σ), the normalized reflected impedanceΔZ_(r)′ can be replaced by ΔZ_(r)′ L/(L+L_(σ)). Likewise, the correctionfactor L/(L+L_(σ)) can be applied to the normalized reflected admittanceΔY_(r)′. As a consequence, the inductance ratio n_(L)=L/L_(p) can bereplaced by (L+L_(σ))/L_(m).

To analyze the circuit 520 with respect to the series resonantconfiguration of the circuit 520 of FIG. 5B, the following assumptionsin addition to the assumption of Equations (23) and (25) are made:

L_(σ)<<L  (90)

ωL_(m)>>R_(Lm)  (91)

n _(T) ² ωL _(m) =α|Z _(11,0)|  (92)

α><1  (93)

The ratio n_(T):1 refers to the transformation ratio of the idealtransformer as used in the transformer's 526 equivalent circuit modelwith reference to FIG. 5H. The factor α determines the impact of thetransformer's main inductance L_(m) on the measured impedance Z_(11,0)and hence on the angle arg{ΔZ} as relevant for purposes of objectdiscrimination as previously discussed with reference to FIG. 5A. Thefactor a is referred herein as to the transformer impact factor. Thelarger α, the less is the impact from the transformer 526. Further,defining a Q-factor Q_(w) for the transformer 526 with respect to itsequivalent series resistance R_(w) (e.g., representing conductorlosses):

Q _(w) ≈ωL _(m) /R _(w)  (94)

and using Equations (30) and (34), the impedance magnitude |Z_(11,0)| atthe measurement port 528 for the series resonant configuration of thecircuit 520 of FIG. 5B may be expressed as:

|Z _(11,0) |≈R _(s) ≈n _(T) ²(R+R _(w))≈n _(T) ²((ω_(s) L/Q)+(ω_(s) L_(m) /Q _(w)))≈n _(T) ²ω_(s) L _(m)/α  (95)

yielding for the inductance ratio n_(L) for satisfying Equation (92):

n _(L) =L/L _(m)≈(Q/α)−(Q/Q _(w))>0  (96)

Equation (95) may also be written as:

|Z _(11,0) |≈R _(s) ≈n _(T) ²(1+Q/(n _(L) Q _(w)))R≈n _(T) ²(Q _(w)/(Q_(w)−α))R  (97)

For Q_(w)>>α and n_(T)>1, the series resonant resistance R_(s) may ben_(T) ² R.

Defining the Q-factor of the series resonant configuration of the sensecircuit 521 as:

Q _(s) ≈n _(T) ²ω_(s) L/R _(s)≈ω_(s) L/(R+R _(w))  (98)

and substituting R_(s) by Equation (97) yields for the Q-factor of theseries resonant configuration of the sense circuit 521 of FIG. 5B:

Q _(s) ≈Q(1−α/Q _(w))  (99)

and for the fractional change using Equation (35):

ΔZ′≈Q _(s) ΔZ _(r) ′≈Q(1−α/Q _(w))ΔZ _(r)′  (100)

The factor 1−α/Q_(w) and thus Q_(s) degrades as Q_(w) decreases or aincreases. This factor may also apply to the SNRs that are related toQ_(s) as given above e.g., by Equations (68) and (85). In an exampleimplementation configured with α=10 and Q_(w)=30, this factor may be ⅔.It may be appreciated that the Q-factor Q_(w) relates to the componentvolume rather than to the transformation ratio n_(T):1.

In some implementations, the transformer impact factor α represents atrade-off between an error in the measured impedance change ΔZ (e.g.,with respect to the angle arg{ΔZ_(r)} as previously discussed withreference to FIG. 5A) and a degradation of the fractional change |ΔZ′|.

To analyze the circuit 520 with respect to the parallel resonantconfiguration of the circuit 520 of FIG. 5B, the following additionalassumption is made:

ωL_(m)>>R_(Lm)  (101)

Defining the inductance ratio:

n _(L) ≈L/L _(m)  (102)

and the Q-factor of the transformer 526 with respect to R_(Lm) (e.g.,core losses) as:

Q_(Lm) ≈ωL _(m) /R _(Lm)  (103)

the admittance |Y_(11,0)| at the measurement port 528 for the parallelresonant configuration of the circuit 520 of FIG. 5B may be expressedas:

|Y _(11,0) ≈G _(p)≈(n _(L) /n _(T) ²)((Q/Q _(w))+(Q/Q _(Lm))+n _(L))/(Qω_(p) L)  (104)

with G_(p) denoting the parallel resonant conductance of the sensecircuit 521. For Q≈Q_(w)≈Q_(Lm) and n_(L)>1, the parallel resonantconductance becomes:

G _(p)≈(n _(L) /n _(T) ²)(n _(L)+2)/(Qω _(p) L)  (105)

Applying Equation (104) to Equation (51), the Q-factor for the parallelresonant configuration of the sense circuit 521 may be expressed as:

Q _(p)≈ω_(p)(L+L _(m))/(R+R _(W) +R _(Lm))≈Q(1+n _(L))/(n _(L)+(Q/Q_(w))+(Q/Q _(Lm)))  (106)

and the fractional change using Equation (53):

ΔY′≈Q _(p) ΔZ _(r) ′n _(L)/(1+n _(L)(≈QΔZ _(r) ′n _(L)/(n _(L)+(Q/Q_(w))+(Q/Q _(Lm)))  (107)

As n_(L) increases, the factor Q_(p)n_(L)f(1+n_(L)) approaches theQ-factor Q of the sense coil 502. In an example implementationconfigured with L_(m)=L (n_(L)=1) and Q_(w)=Q_(Lm)=Q, this factor may beQ/3. For n_(L)>>1, the fractional change ΔY′ may equal ΔZ′ of the seriesresonant configuration of the circuit 520 of FIG. 5B as given byEquation (100). For the example parallel resonant configuration asspecified above and for an example series resonant configuration withα/Q_(w)=⅓, equality may occur at n_(L)=4.

Based on Equation (107), an example implementation of the circuit 520configured for parallel resonance with L_(m)=L (n_(Lm)=1) andQ_(w)=Q_(Lm)=Q and a transformer 526 with a transformation ratio n_(T)²≈⅓ provides a fractional change ΔY′≈0.33Q ΔZ_(r)′. Another exampleimplementation of the circuit 520 using a transformer 526 with n_(T)=1and Q_(w)=Q_(Lm)=Q and an inductance ratio n_(L)≈2.16 to provide thesame admittance |Y₁₁|, yields a fractional change ΔY′≈0.52Q ΔZ_(r)′. Afurther example implementation of the circuit 500 (without transformer526) with Q_(Lp)=Q with an inductance ratio n_(L)=1 and yields afractional change ΔY′≈0.72Q ΔZ_(r)′.

From above examples, it may be concluded that using the transformerlesscircuit 500 may be preferable in a parallel resonant configuration. If atransformer (e.g., transformer 526) is indispensable e.g., for purposesof balancing as previously discussed with reference to FIG. 5B,decreasing the inductance ratio n_(L) rather than n_(T) may result in alarger fractional change.

The equivalent circuit model 540-1 as illustrated in FIG. 5G comprisesthe sense coil's 502 inductance L and its equivalent parallelconductance G, the parallel capacitor's 544 capacitance C_(p), and theseries capacitor's 546 capacitance C an ideal sense signal voltagesource 552, and an ideal current measurement circuit 550. It may beappreciated that in practical implementations, losses in the capacitorsare generally substantially lower than losses in inductors. Therefore,the equivalent series resistance of the capacitors 544 and 546 areneglected (not shown) in the equivalent circuit model 540-1 of FIG. 5G.Further, the equivalent circuit model 540-1 includes an admittanceΔY_(r) in parallel to the inductance L representing the reflectedadmittance of the object 110, 112, or 114 proximate to the sense coil502. (The reflected admittance ΔY_(r) may be regarded as the object 110,112, or 114 as illustrated in FIG. 5C abstracted away). The equivalentcircuit model 540-1 also includes a noise current source I_(sn) inparallel to the inductance L representing the noise current inductivelyand capacitively coupled into the sense coil 502 by the magnetic andelectric field, respectively, as generated when WPT is active. The noisecurrent I_(sn) may include any low frequency component (e.g., thefundamental of the WPT operating frequency and harmonics thereof) aswell as any high frequency component (e.g., switching noise at the sensefrequency). The equivalent circuit model 540-1 further indicates theadmittance Y₁₁ and the impedance Z₁₁ (=1/Y₁₁), the sense signal sourcevoltage V₀ with an additive noise voltage component V_(0,n), the sensesignal current I with an additive noise current component I_(n), and themeasurement port 548 (indicated by the terminal and the dashed line)where the voltage V₀+V_(0,n) is applied, the current I+I_(n) ismeasured, and where Y₁₁ or Z₁₁ refer to. Because the equivalent circuitmodel 540-1 applies to the circuit 540 of FIG. 5C, the reference numeral540 is used instead in the following theoretical analysis.

With the assumption of an identical sense coil 502 in the circuits 500and 540, the following relations may apply:

ΔY_(r)′=ΔZ_(r)′  (108)

G≈R/(ωL)²  (109)

ΔY _(r) ≈ΔZ _(r)/(ωL)²  (110)

I _(sn) ≈V _(sn)/(ωL)  (111)

with ΔY_(r)′, ΔZ_(r), R, and V_(sn) referring to the normalizedreflected admittance, the normalized reflected impedance, the reflectedimpedance of the object 110 in the sense coil 502, the equivalent seriesresistance of the sense coil 502, and the disturbance voltage V_(sn)with reference to the circuit 500 of FIG. 5F, respectively.

To analyze the series and parallel resonant configuration of the circuit540 of FIG. 5G, the common assumptions:

1/ωL>>G  (112)

|ΔY _(r) |<<G  (113)

are made for a frequency range about the resonant frequency.

In an implementation configured for parallel resonance and with asusceptance:

ωC _(s) >>|Y ₁₁|  (114)

in a frequency range about the resonant frequency, the admittance Y₁₁ atthe measurement port 548 of the circuit 540 of FIG. 5G in presence of anobject (e.g., object 110) may be expressed as:

Y ₁₁ ≈G+(jωL)⁻¹ +jωC _(p) +ΔY _(r)  (115)

In absence of a foreign object, a local minimum of |Y_(11,0)(ω)|(parallel resonance) occurs substantially at an angular frequency ωsatisfying:

(jωL)⁻¹ +jωC _(p)≈0  (116)

yielding the parallel resonant angular frequency:

ω_(p)≈(L C _(p))^(−1/2)  (117)

At this frequency, the admittance Y_(11,0) becomes approximately real:

Y _(11,0) ≈Re{Y _(11,0) }=G _(p) ≈G  (118)

with G_(p) denoting the parallel resonant conductance, while theadmittance Y₁₁ in presence of an object (e.g., object 110) isapproximately:

Y ₁₁ ≈G _(p) +ΔY≈G+ΔY _(r)  (119)

with ΔY_(r) referring to the reflected admittance as previously definedwith reference to FIG. 5A.

Applying Equations (118) and (119) to Equation (9), the fractionalchange ΔY′ for the parallel resonant configuration of the circuit 540 ofFIG. 5G becomes approximately:

ΔY′≈ΔY/G _(p) ≈ΔY _(r) /G  (120)

Defining the normalized reflected admittance:

ΔY_(r)′=ΔY_(r)ω_(p)L  (121)

the Q-factor of the sense coil 502:

Q=1/(ω_(p) LG)  (122)

and the Q-factor of the parallel resonant configuration of the sensecircuit 541 of FIG. 5G:

Q _(p)≈1/(ω_(p) LG _(p))≈Q  (123)

the fractional change may also be written in terms of ΔY_(r)′ and Q_(p):

ΔY′≈Q_(p)ΔY_(r)′  (124)

To analyze the series resonant configuration of the circuit 540 of FIG.5G, the additional assumption:

|ωC _(p)−(ωL)⁻¹ |>>G  (125)

is made for a frequency range about the resonant frequency. Theimpedance Z₁₁ at the measurement port 548 in presence of an object(e.g., object 110) may be expressed as:

Z ₁₁=(jωC _(s))⁻¹+(G+jωC _(p)+(jωL)⁻¹ +ΔY _(r))⁻¹  (126)

Using Equations (112), (113), (125), and (38), Equation (126) may beapproximated as:

Z ₁₁=(jωC _(s))⁻¹+(jωC _(p)+(jωL)⁻¹)⁻¹)+(G+ΔY _(r))(ωC_(p)+(ωL)⁻¹)⁻²  (127)

In absence of a foreign object, a local minimum of |Z_(11,0)(ω)| (seriesresonance) occurs substantially at an angular frequency ω satisfying:

(jωL)⁻¹ +jω(C _(p) +C _(s))≈0  (128)

yielding for the series resonant angular frequency:

ω_(s)≈(L(C _(p) +C _(s)))^(−1/2)  (129)

At this frequency, the impedance Z_(11,0) becomes substantially real:

Z _(11,0) ≈Re{Z _(11,0) }R _(s) ≈G/(ω_(s) C _(s))²  (130)

with R_(s) denoting the series resonant resistance, while the impedanceZ₁₁ in presence of an object (e.g., object 110) is approximately:

Z ₁₁ ≈R _(s) +ΔZ≈(G+ΔY _(r))/(ω_(s) C _(s))²  (131)

with:

ΔZ≈ΔY _(r)/(ω_(s) C _(s))²  (132)

Further, defining the Q-factor of the sense coil 502:

Q=1/(ω_(s) LG)  (133)

and the capacitance ratio:

n _(C) =C _(p) /C _(s)  (134)

the impedance Z_(11,0) of Equation (130) at ω_(s) may be expressed as:

Z _(11,0) =R _(s)≈1/(Qω _(s) Lω _(s) ² C _(s) ²)≈(1+n _(C))²ω_(s)L/Q  (135)

For n_(C)>>1, the series resonant resistance R_(s) becomesapproximately:

R _(s) ≈n _(C) ²ω_(s) L/Q  (136)

and approximately 9 R for the case n_(C)=2. According to Equation (135),the impedance Z₁₁ at ω_(s) of the sense circuit 541 can be modified(e.g., increased) by adjusting the capacitance ratio n_(C)=C_(p)/C,accordingly, while maintaining series resonance substantially at thenominal sense frequency. Therefore, in some implementations, the seriesresonant configuration of the circuit 540 of FIG. 5G is employed as analternative to using a transformer (e.g., transformer 526 of FIG. 53)for transforming the impedance Z₁₁ to be within a suitable operatingimpedance range as previously discussed with reference to FIG. 5B.

Applying Equations (130) and (131) to Equation (8), the fractionalchange ΔZ′ for the series resonant configuration of the circuit 540 ofFIG. 5G becomes approximately:

ΔZ′=ΔZ/R _(s) ≈ΔY _(r) /G  (137)

showing that the impedance change ΔZ is substantially proportional tothe reflected admittance ΔY_(r). Therefore, the angle arg{ΔZ} of themeasured impedance change ΔZ may be indicative of the angle arg{ΔY_(r)}.As previously described with reference to the circuit 540 of FIG. 5C,the accuracy of the measured angle may be improved by applyingcalibration.

Defining the Q-factor of the series resonant configuration of the sensecircuit 541 of FIG. 5G:

Q _(s)≈ω_(s)(C _(s) +C _(p))/G≈Q  (138)

which approximately equals the Q-factor Q of the sense coil 502 asdefined by Equation (133), using Equation (7), and applying Equation(133) to (137), the fractional impedance change ΔZ′ may also be writtenas:

ΔZ≈Q _(s) ΔY _(r) ′≈QΔY _(r)′  (139)

In a further aspect, the drive voltage level V₀, the resulting current Iat the measurement port 548, and the drive power level P are considered.In some implementations based on the circuit 540 of FIG. 5G, the voltagelevel V₀ of the voltage source 542 is adjusted to achieve a specifiedcurrent level |I_(L)| in the sense coil 502. For the parallel resonantconfiguration of the circuit 540 of FIG. 5G, the voltage level V₀approximately equals the voltage across the sense coil 502 providing therelation:

V ₀≈ω_(p) L|I _(L)|  (140)

Using Equations (118) and (122), the current I at the measurement port548 may be expressed approximately as:

I≈|Y _(11,0) |V ₀ ≈|I _(L)|ω_(p) LG=|I _(L) |/Q  (141)

and the drive power level:

P≈V ₀ I=|I _(L)|²ω_(p) L/Q  (142)

For the series resonant configuration of the circuit 540 of FIG. 5G, thevoltage |V_(L)| across the sense coil 502 is approximately:

|V _(L)|≈ω_(s) L|I _(L)|  (143)

and the current I using Equations (129), (134), and (143):

I≈|V _(L)|ω_(s) C _(s)=ω_(s) ² LC _(s) |I _(L) |=|I _(L) |C _(s)/(C _(s)+C _(p))=|I _(L)|/(1+n _(C))  (144)

respectively. The voltage V₀ and the drive power P required to drive thesense coil 502 with a current level |I_(L)| can be found to be:

V ₀ ≈IR _(s) ≈I(1+n _(C))R=|I _(L)|ω_(s) L/Q _(s)  (145)

P=I ² R _(s) ≈|I _(L)|²ω_(s) L/(Q _(s)(1+n _(C)))  (146)

In a further aspect, it may be meaningful to define the narrowband SNRat the measurement port 548 of the circuit 540 of FIG. 5G as given byEquation (14), where |ΔI| denotes the magnitude of the current change inthe measured current I due to the presence of an object (e.g., object110) and I_(n) the additive noise voltage component as indicated in thecircuit 540 of FIG. 5G. More specifically, the current change |ΔI| mayrefer to the r.m.s. current and In to the r.m.s. noise current asmeasured at the nominal sense frequency in the bandwidth B_(m) of thecurrent measurement circuit 550. This noise current I_(n) may includecircuit intrinsic and extrinsic noise components as discussed above. TheSNR as given by Equation (14) is referred herein as to the differentialnarrowband SNR.

In another aspect, it may be meaningful to define the broadbandextrinsic SNR at the measurement port 548 of the circuit 540 of FIG. 5Gas:

SNR _(W) =|I|/I _(W)  (147)

where |I| denotes the magnitude of the sense signal current and I_(w)the disturbance current at the fundamental WPT operating frequency,which may be a prominent component in I_(in) when WPT is active. Morespecifically, the current |I| may refer to the r.m.s. current and I_(w)to the r.m.s. disturbance current as measured at the measurement port548 at the fundamental WPT operating frequency.

Using Equation (14) and (7), the differential narrowband extrinsic SNRof the parallel resonant configuration of the circuit 540 of FIG. 5G maybe expressed as:

ΔSNR _(ex,p) ≈|ΔY _(r)|ω_(p) L|I _(L) |/I _(sn) =|ΔY _(r) ′||I _(L) |I_(sn)  (148)

with I_(sn) the noise current as illustrated in FIG. 5G.

Since the sense circuit 541 transforms the shunt current through ΔY_(r)to ΔI in the same way as it transforms I_(sn) to I_(n), Equation (148)also applies to the series resonant configuration, meaning that:

ΔSNR_(ex,s)≈ΔSNR_(ex,p)  (149)

In operations of the circuit 540 where the noise voltage V_(0,n) ispredominant as previously discussed, the noise current I_(n) in theparallel resonant configuration of the circuit 540 is approximately:

I_(n)≈G_(p)V_(0,n)  (150)

and the current change in presence of an object (e.g., object 110) is:

|ΔI|≈|V ₀ ||ΔY _(r)|  (151)

Applying Equations (123), (124), (150), and (151) to Equation (14), thedifferential narrowband intrinsic SNR with respect to the noise voltageV_(0,n) for the parallel resonant configuration of the circuit 540 ofFIG. 5G may be expressed as:

ΔSNR _(int,p)≈(|V ₀ |/V _(0,n))|ΔY _(r) |/G _(p)≈(|V ₀ |/V_(0,n))|ΔY′|  (152)

Using Equation (124), Equation (152) may also be written in terms of theQ-factor Q_(p) and the normalized reflected admittance ΔY_(r)′ as:

ΔSNR _(int,p)≈(|V ₀ |/V _(0,n))Q _(p) |ΔY _(r)′|  (153)

Using:

ΔZ′<<1  (154)

which follows from assumptions of Equations (112) and (113) and Equation(38), the magnitude |ΔI| of the current change in the series resonantconfiguration of the circuit 540 of FIG. 5G may be approximated as:

|ΔI|=|(V ₀ /Z ₁₁)−(V ₀ /Z _(11,0))=|V ₀||(Z _(11,0) +ΔZ)⁻¹ −Z _(11,0))⁻¹|≈|V ₀ ||ΔZ|/|Z _(11,0)|²  (155)

With the noise voltage V_(0,n) as the predominant noise contribution,the noise current I_(n) at the series resonant frequency becomes:

I _(n) =V _(0,n) /|Z _(11,0)|  (156)

Applying Equations (155), (156), and (137) to Equation (14), thedifferential narrowband intrinsic SNR with respect to the noise voltageV_(0,n) for the series resonant configuration of the circuit 540 of FIG.5G may be expressed as:

ΔSNR _(int,s)≈(|V ₀ |/V _(0,n))|ΔZ′|≈(|V ₀ |/V _(0,n))|ΔY _(r)|/G  (157)

Using Equation (139), Equation (157) may also be written in terms of theQ-factor Q_(s) and the normalized reflected admittance ΔY_(r)′ as:

ΔSNR _(int,s)≈(|V ₀ |/V _(0,n))Q _(s) |ΔY _(r)′|  (158)

Similar considerations may be made for thermal noise, though likely lesssignificant in practical implementations as shown below with referenceto TABLE 2. As previously mentioned, a thermal noise current isgenerated by the equivalent parallel conductance G of the sense coil502. The noise current I_(n) may be considered as the thermallygenerated by the parallel resonant conductance G_(p) as defined byEquation (118) and becomes approximately:

I _(n)=(4kTB _(m) G _(p))^(1/2)≈(4kTB _(m) G)^(1/2)  (159

Applying Equation (151), (159), and (7) to Equation (14), thedifferential narrowband intrinsic SNR with respect to thermal noise ofthe parallel resonant configuration of the circuit 540 may be expressedas:

ΔSNR _(int,p) ≈|I _(L)|ω_(p) L|ΔY _(r) |/I _(n) ≈|I _(L) ||ΔY_(r)′|/(4kTB _(m) G)^(1/2)  (160)

where k denotes the Boltzmann constant, T the absolute temperature, andB_(m) the equivalent noise bandwidth of the current measurement circuit550.

In the series resonant configuration of the circuit 540, the noisecurrent component I_(n) as thermally generated by the series resonantresistance R_(s) as defined by Equation (118) becomes:

I _(n)=(4kTB _(m) /R _(s))^(1/2)  (160)

Using Equations (137), (129), and the relation:

|V ₀ |=|I|R _(s) ≈|I _(L)|ω_(s) ² LC _(s) R _(s)  (162)

the current change |ΔI| may be expressed as:

|ΔI|≈|V ₀ ||ΔZ|/R _(s) ² ≈|I _(L)|ω_(s) ² LC _(s) |ΔZ′|≈|I _(L) ||ΔY_(r)|/((1+n _(C))G)  (163)

Applying Equation (161) and (163) to Equation (14), the differentialnarrowband intrinsic SNR with respect to thermal noise of the seriesresonant configuration of the circuit 540 may be expressed in terms ofthe sense coil 502 current IILI and the normalized reflected admittance|ΔY_(r)′| as:

ΔSNR _(int,s) ≈|L _(L) ||ΔY _(r)′|/(4kTB _(m) G)^(1/2)  (164)

In yet a further aspect, the broadband extrinsic SNR as defined byEquation (147) with respect to the induced current component I_(SW) atthe fundamental WPT operating angular frequency ω_(W) is considered.Assuming:

I_(sn)=I_(sW)  (165)

1/(ω_(W) C _(p))>>ω_(W) L  (166)

and using Equation (117), the disturbance current component I_(W) in thecurrent I for the parallel resonant configuration of the circuit 540 ofFIG. 5G becomes approximately:

I _(n) =I _(W) ≈I _(sW)ω_(W) Lω _(W) C _(s) ≈I _(sW)(ω_(W)/ω_(p))² /n_(C)  (167)

The factor (ω_(W)/ω_(p))²/n_(C) may be considered as the attenuation ofthe low frequency induced current I_(sW) by the high pass filter effectof the sense circuit 541. Using:

|I|≈|V _(L) |G  (168)

and applying Equations (167), (168), and (134) to Equation (147), thebroadband extrinsic SNR of the parallel resonant configuration of thecircuit 540 of FIG. 5G may be expressed as:

SNR _(W,p)≈(|V _(L) |G/I _(sW))(ω_(p)/ω_(W))² n _(C)  (169)

with V_(L) denoting the voltage across the sense electrode 702.

Using the relation:

|V _(L) |≈|I _(L)|ω_(p) L  (170)

and Equation (123), Equation (169) may also be written as:

SNR _(W,p)≈(|I _(L) |/I _(sW))(ω_(p)/ω_(W))² n _(C) /Q _(p)  (171)

Using Equations (82), (129), (134), and (166), the disturbance currentI_(w) in the current I for the series resonant configuration of thecircuit 540 of FIG. 5G becomes approximately:

I _(W) ≈I _(sW)ω_(W) Lω _(W) C _(s) ≈I _(sW)(ω_(W)/ω_(s))²(1+n_(C))  (172)

The factor (ω_(W)/ω_(p))²(1+n_(C)) may be considered as the attenuationof the low frequency induced current I_(sw) by the high pass filtereffect of the sense circuit 541. Further, expressing the sense signalcurrent |I| at the angular frequency co, in terms of the sense coil 502voltage |V_(L)| using Equations (138) and (168):

|I|≈|V _(L)|(ω_(s) C _(s)−(ω_(s) L)⁻¹)≈|V _(l)|ω_(s) C _(s)  (173)

and applying Equations (172) (173), and (134) to Equation (14), thebroadband extrinsic SNR with respect to the WPT fundamental disturbancecurrent component I_(sW) for the series resonant configuration of thecircuit 540 of FIG. 5G may be expressed as:

SNR_(W.s)≈(|V _(L)|ω_(s) C _(s) /I _(sW))(ω_(s)/ω_(W))²(1+n _(C))  (174)

Using the relation:

|V _(L) |≈|I _(L)|ω_(s) L  (175)

Equations (129), and (165), Equation (174) may also be written as:

SNR _(W,s)≈(|I _(L) |/I _(sW))(ω_(s)/ω_(W))²  (176)

Based on Equations (171) and (176) and ω_(s)=ω_(p), the followingrelation between the broadband extrinsic SNRs of the parallel and seriesresonant configurations of the circuit 540 of FIG. 5G can be found:

SNR _(W,s) ≈SNR _(W,p) Q _(p) /n _(C)  (177)

TABLE 1 provides example parameter values as used for a numericalanalysis of the series and parallel resonant configuration of thecircuit 500 of FIG. 5F and the circuit 540 of FIG. 5G. Values for theinduced disturbance voltage V_(sW), the noise voltage V_(sn), and theirequivalent respective currents I_(sW) and I_(sn) of the circuit 540 maybe considered typical for the multi-purpose detection circuit 100integrated into a wireless power transfer structure (e.g., wirelesspower transfer structure 200 with reference to FIG. 2). The normalizedreflected impedance of the object 110 as given in TABLE 1 may be typicalfor a paperclip placed on the surface of the wireless power transferstructure 200 at a worst case position 3 mm above the sense coil 502(e.g., inductive sense element 107 a) with a form factor of 60×80 mm.The example sense current level |I_(L)| may be within a constraint givenby an electromagnetic emission limit of an established electromagneticcompatibility (EMC) standard (e.g., EN 300330). The example drive signalSNR |I₀|/I_(0,n) and |V₀|/V_(0,n) for the circuit 500 and 540,respectively, may be typical for a digital implementation of a sensesignal source (e.g., sense signal current source 512 and sense signalvoltage source 552), respectively, as previously described withreference to FIG. 4.

TABLE 1 Circuit 500 of FIG. 5F 540 of FIG. 5G Series Parallel SeriesParallel Configuration resonant resonant resonant resonant Nominal sense3 MHz 3 MHz 3 MHz MHz frequency WPT operating 85 kHz 85 kHz 85 kHz 85kHz frequency Inductance L of sense 5 μH 5 μH 5 μH 5 μH coil 502Inductance/capacitance n_(L) = 1 n_(L) = 2.5 n_(C) = 2 n_(C) = 1 ratioQ-factor Q of sense coil 30 30 30 30 502 Q-factor of capacitorsQ_(Cs) >> Q Q_(Cs) >> Q Q_(Cp) >> Q Q_(Cp) >> Q 504/544 Q-factor ofinductor Q_(Lp) = Q Q_(Lp) = Q Q_(Cs) >> Q Q_(Cs) >> Q 506/capacitor 546Normalized reflected |ΔZ_(r)′| = |ΔZ_(r)′| = |ΔY_(r)′| = |ΔY_(r)′| =impedance/admittance 100 ppm 100 ppm 100 ppm 100 ppm Angle of reflectedarg{ΔZ_(r)} = arg{ΔZ_(r)} = arg{ΔY_(r)} = arg{ΔY_(r)} =impedance/admittance 45° 45° 45° 45° Sense coil current level 20mA_(rms) 20 mA_(rms) 20 mA_(rms) 20 mA_(rms) |I_(L)| Extrinsic noisevoltage 10 μV_(rms) 10 μV_(rms) 0.11 μA_(rms) 0.11 μA_(rms)V_(sn)/current I_(sn) (WPT switching noise) SNR of sense signal|I₀|/I_(0, n) = |I₀|/I_(0, n) = |V₀|/V_(0, n) = |V₀|/V_(0, n) = source512/552 80 dB 80 dB 80 dB 80 dB Ambient temperature T 350 K 350 K 350 K350 K Equiv. noise bandwidth 200 Hz 200 Hz 200 Hz 200 Hz B_(m) ofmeasurement circuit 510/540 WPT fundamental 30 V_(rms) 30 V_(rms) 11.2A_(rms) 11.2 A_(rms) disturbance voltage V_(sW)/current I_(sW)

Numerical results as obtained from a circuit analysis using thenumerical assumptions of TABLE 1 are listed in TABLE 2. While the SNRvalues are obtained using the corresponding approximate equations asdefined above with reference to FIGS. 5F and 5G, the values related toinductances, capacitances, impedances, fractional change, currents,voltages, and power result from using a more accurate analytical tool.TABLE 2 also includes numerical results for the angle error in themeasured impedance change ΔZ e.g., in presence of the object 110. In amultipurpose detection circuit 100 employing an angle calibrationprocedure as previously described with reference to FIGS. 5A. For theseries resonant configuration of the circuit 500 of FIG. 5F, the angleerror is defined as:

ε≈arg{ΔZ exp(−j arg{Z _(11,0)})}−arg{ΔZ _(r)′}  (178)

where ΔZ exp(−j arg{Z_(11,0)}) denotes the impedance change with theangle correction applied. For the parallel resonant configuration of thecircuit 500 of FIG. 5F, it is defined as:

ε≈arg{ΔY exp(−j arg{Y _(11,0)})}−arg{ΔZ _(r)′}  (179)

For the parallel resonant configuration of the circuit 540 of FIG. 5G,the angle error is defined as:

ε≈arg{ΔY exp(−j arg{Y _(11,0)})}−arg{ΔZ _(r)′}  (180)

and for the series resonant configuration of the circuit 540 of FIG. 5G,it is defined as:

ε≈arg{ΔZ exp(−j arg{Z _(11,0)})}−arg{ΔZ _(r)′}  (181)

Further, TABLE 2 includes the drive current level I₀, the drive powerlevel P required to drive the sense coil 502 of the sense circuit 501with the sense current |I_(L)| as specified in TABLE 1. Accordingly, itincludes the drive voltage level V₀, the drive power level P required todrive the sense coil 502 of the sense circuit 541 with the sense current|I_(L)| as specified in TABLE 1.

TABLE 2 Circuit 500 of FIG. 5F 540 of FIG. 5G Series Parallel SeriesParallel Configuration resonant resonant resonant resonant Capacitanceof capacitor C_(s) = 563 pF C_(s) = 402 pF C_(p) = 375 pF C_(p) = 563 pF504/544 Inductance/capacitance of L_(p) = 5 μH L_(p) = 2 μH C_(s) = 188pF C_(s) = 563 pF inductor/capacitor 506/546 Q-factor of sense circuitQ_(s) ≈ 30 Q_(p) ≈ 30 Q_(s) ≈ 30 Q_(p) ≈ 30 501/541 Precise frequency of3.0017 MHz 2.9959 MHz 2.9967 MHz 3.0017 MHz minimum|Z_(11.0)|/|Y_(11.0)| Impedance |Z_(11.0)| of sense 3.13 Ω 326 Ω 28.1 Ω2.8 kΩ circuit 501/541 Fractional change |ΔZ′| 0.30% 0.21% 0.30% 0.30%Impedance angle error ε −0.04°  −2.0°   −1.9°   −0.04°  Drive currentI₀/ ≈20 mA_(rms) ≈2.3 mA_(rms) ≈0.19 V_(rms) ≈1.9 V_(rms) voltage V₀Voltage across |Z_(11.0)|/ ≈63 mV_(rms) ≈0.76 V_(rms) ≈6.7 mA_(rms)≈0.67 mA_(rms) current through |Y_(11.0)| Drive power P ≈1.3 mW 1.8 mW≈1.3 mW ≈1.3 mW Differential narrow-band ΔSNR_(ex, s) ≈ 25.5ΔSNR_(ex, p) ≈ 25.5 ΔSNR_(ex, s) ≈ 25.5 ΔSNR_(ex, p) ≈ 25.5 extrinsicSNR (WPT switching dB dB dB dB noise) Differential narrow-bandΔNR_(int, s) ≈ 29.5 ΔSNR_(int, p) ≈ 26.6 ΔSNR_(int, s) ≈ 29.5ΔSNR_(int, p) ≈ 29.5 intrinsic SNR (Sense signal dB dB dB dB noise)Differential narrow-band ΔSNR_(int, s) ≈ 94.7 ΔSNR_(int, p) ≈ 93.2ΔSNR_(int, s) ≈ 94.7 ΔSNR_(int, p) ≈ 94.7 intrinsic SNR (Thermal noise)dB dB dB dB Broadband extrinsic SNR SNR_(W, s) ≈ 8.3 SNR_(W, p) ≈ 40.8SNR_(W, s) ≈ 6.9 SNR_(W, p) ≈ −22.6 (WPT fundamental dB dB dB dBdisturbance)

Based on the numerical results of TABLE 2, the following conclusions maybe drawn. The high impedance magnitude |Z_(11,0)| as generally presentedby the parallel resonant configuration of the circuit 500 of FIG. 5A canbe substantially decreased with a moderate loss in fractional change byconfiguring the sense circuit 501 with an inductance ratio n_(L)>1(e.g., n_(L)=2.5). Conversely, the low impedance magnitude |Z_(11,0)| asgenerally presented by the series resonant configuration of the circuit540 of FIG. 5C can be increased without loss in fractional change byconfiguring the sense circuit 541 with a capacitance ratio n_(C)>1(e.g., n_(C)=2). Further, the results in TABLE 2 show circuits andconfigurations equivalent in terms of the differential narrowbandextrinsic SNR (WPT switching noise). Moreover, the numbers for thedifferential narrowband intrinsic SNR (sense signal noise) show theparallel resonant configuration of the circuit 500 slightly inferior tothe other circuits and configurations. The high numbers obtained for thedifferential narrowband intrinsic SNR (thermal noise) show that thermalnoise is negligible, even when WPT is inactive. The numbers resultingfor the broadband extrinsic SNR (WPT fundamental disturbance) show asubstantial difference (>60 dB) between the parallel resonantconfiguration of the circuit 500 and 540. The series resonantconfiguration of the circuit 500 and 540 are almost equivalent and theSNRs are slightly above 6 dB, which may be a minimum requirement in apractical implementation. TABLE 2 further shows a negligible angle error|ε| for the series resonant configuration of the circuit 500 and theparallel resonant configuration of the circuit 540 and an angle error ofabout 2° for each of the other configurations. Finally, the current orvoltage levels as required at the respective measurement port 508 and548 for driving the sense coil 502 with the specified sense currentlevel |I_(L)| may be within suitable ranges of low power electronics forthe circuits and configurations as theoretically analyzed herein.

FIG. 5H illustrates an “L” equivalent circuit model 526-1 applicable tothe non-ideal transformer 526 and 726 with reference to the circuit 520of FIG. 5B and the circuit 720 of FIG. 7C, respectively. The “L”equivalent circuit comprises an ideal transformer (indicated by theinfinity symbol) with transformation ratio n_(T):1, a secondary referredmain inductance L_(m) , and a secondary referred series (leakage)inductance L_(σ).

FIG. 5I illustrates a “T”-equivalent circuit model 562-1 applicable tothe double-coil inductive sense elements 562 used in the circuit 560 ofFIG. 5D and the circuit 580 of FIG. 5E. The circuit model 562-1comprises three inductances connected in a “T”-topology and related tothe inductance L₁, L₂, and the mutual inductance L_(M) as indicated inFIGS. 5D and 5E.

FIG. 5J illustrates another equivalent circuit model 562-2 applicable tothe double-coil inductive sense element used in the circuit 560 of FIG.5D and the circuit 580 of FIG. 5E. The circuit model 562-2 comprises theinductances L₁ and L₂ in series to the respective current-controlledvoltage sources V_(ind,1) and V_(ind,2) representing the voltage inducedinto the first and second sense coil, respectively.

FIG. 5K shows a table of a summary of selected equations with respect tothe resonant frequency, the Q-factor of the sense circuit, theimpedance/admittance of the sense circuit, the fractional change, andthe various SNRs for the series and parallel resonant configurations ofthe circuit 500 of FIG. 5F and the circuit 540 of FIG. 5G. As previouslynoted, these equations are valid for the assumptions made with referenceto FIGS. 5F and 5G.

FIG. 6 illustrates a complex plane 600 or more precisely a complex halfplane comprising quadrant 1 and 4 where the reflected impedances ΔZ_(r)of different types (categories) of objects (e.g., object 110, 112, 114,or vehicle 330) may occur if proximate to a sense coil (e.g., sense coil502 with reference to FIG. 5A). More specifically, FIG. 6 shows shadedareas (e.g., angle ranges 602 to 610) where the reflected impedancesΔZ_(r) of different types (categories) of objects (e.g., object 110,112, 114) may be measured at a sense frequency (e.g., in the MHz range).To emphasize the characteristics of the different categories of objects,the angle ranges 602 to 610 indicated in FIG. 6 may be not drawn toscale and should be considered qualitative rather than quantitative. Theactual angle ranges may also depend on the particular sense frequency,certain characteristics of the inductive sense element (e.g., sense coil502), the capacitive sensing effect of the inductive sense element aspreviously discussed with reference to FIG. 1, the position andorientation of an object relative to the inductive sense element.

The complex plane 600 and the shaded areas (e.g., angle ranges 602 to610) may also apply to the reflected admittance ΔY_(r) by simplyrelabeling the real and imaginary axis by Re{ΔY_(r)} and j Im{ΔY_(r)},respectively (not shown in FIG. 6).

Further, FIG. 6 illustrates different types of metallic objects 110 suchas a 1 € cent coin (object 110 a), a metal foil (object 110 b), a steelnut (object 110 c), a steel nail, a fixing pin, and steel wire pieces(objects 110 d). Moreover, it illustrates different types of non-living,substantially non-conductive or weakly conductive objects 112 such as aferrite core (object 112 a), a plastic bottle filled with water (object112 b), and a living object 114 representing a hand (symbolizing a humanextremity).

The angle range 602 (e.g., close to −90°) in quadrant 4 may becharacteristic for an object (e.g., object 110) exhibiting a relativelyhigh electric conductivity (e.g., σ>50 MS/m) and substantially noferromagnetic effect (relative permeability μ_(r)≈1) at the sensefrequency. For a sense frequency in the MHz range, the impedance changeΔZ caused by a copper coated coin (e.g., object 110 a) may cause animpedance change ΔZ in the angle range 602.

The angle range 604 (e.g., around −80°) in quadrant 4 may becharacteristic for an object (e.g., 110) exhibiting a substantiallylower equivalent conductivity (e.g., σ>5 MS/m) and substantially noferromagnetic effect (relative permeability μ_(r)≈1) at the sensefrequency. A piece of thin foil or metallized paper (e.g., aluminumcoated paper) as illustrated in FIG. 6 by object 110 b (e.g., with athickness of the metal layer smaller than the skin depth δ at the sensefrequency) may reflect an impedance ΔZ_(r) in the angle range 604 for asense frequency in the MHz range.

The angle range 606 (around 0°) in quadrant 4 and 1 may becharacteristic for an object (e.g., object 110) exhibiting a relativelyhigh conductivity (e.g., σ>10 MS/m) and a substantial ferromagneticeffect (e.g., μ_(r)>50) at the sense frequency. An object made offerromagnetic steel (e.g., object 110 c) may reflect an impedance ΔZ_(r)in the angle range 606 for a sense frequency in the MHz range.Ferromagnetism (μ_(r)>1) in the metallic object 110 d generally reflectsan impedance ΔZ_(r) with an imaginary part Im{ΔZ_(r)}>0. On the otherhand, the electrical conductivity of the metallic object 110 d generallyreflects an impedance ΔZ_(r) with Im{ΔZ_(r)}<0 and Re{ΔZ_(r)}>0.Superimposing the two opposing effects may result in a net reflectedimpedance ΔZ_(r) e.g., in the angle range 606.

The angle range 608 (e.g., around 45°) in quadrant 1 may becharacteristic for an object (e.g., object 110) exhibiting a relativelyhigh conductivity (e.g., σ>10 MS/m) and a substantial ferromagneticeffect (e.g., μ_(r)>50) at the sense frequency and with a lengthsubstantially larger than a thickness. An object made of ferromagneticsteel (e.g., one of the objects 110 d) may cause an impedance change ΔZin the angle range 606 for a sense frequency in the MHz range.Ferromagnetism (μ_(r)>1) in the metallic object 110 d generally reflectsan impedance ΔZ_(r) with a positive imaginary part that prevails theconductivity effect acting in the opposite direction as described abovewith reference to the object 110 c. Superimposing the two effectsresults is a net reflected impedance ΔZ_(r) with a positive imaginarypart (Im{ΔZ_(r)}>0) substantially equal to the real part Re{ΔZ_(r)}corresponding to the angle range 608. A reflected impedance ΔZ_(r) inthis angle range or may also be caused by a paper clip made offerromagnetic steel (not shown in FIG. 6).

Finally, the angle range 610 (e.g., close to 90°)in the quadrant 1 maybe characteristic for a substantially non-conductive object (e.g.,object 112) that exhibits a dielectric effect (ε_(r)>1) at the sensefrequency. A dielectric object (e.g., object 112 b) may cause areflected impedance ΔZ_(r) in the angle range 610. A living object(e.g., object 114) may also reflect an impedance ΔZ_(r) in the anglerange 610. As previously discussed in connection with FIG. 5A,dielectric objects (e.g., object 112 or 114) may interact with the sensecoil (e.g., sense coil 502 of FIG. 5A) via the electric stray fieldgenerated by the sense coil's parasitic capacitances (e.g., C_(iw),C_(gnd), and C_(wpt)) as illustrated in FIG. 5A. Further, the anglerange 610 (e.g., close to 90°) in quadrant 1 may be characteristic for asubstantially non-conductive object (e.g., object 112) that exhibits aferromagnetic effect (μ_(r)>1) at the sense frequency. An object made offerrite material (e.g., object 112 a) may reflect an impedance changeΔZ_(r) in the angle range 610.

In an aspect of the multi-purpose detection circuit 100, objects 110(e.g., object 110 a, 110 b, 110 c, 110 d) producing a reflectedimpedance ΔZ_(r) in the respective angle ranges 602, 604, 606, and 608or somewhere between these ranges may be subject of induction heating ifexposed to the strong WPT magnetic field. This may be particularly truefor thin foils (e.g., object 110 b) and objects that are bothsubstantially electrically conductive and ferromagnetic (e.g., objects110 c and 110 d). Ferromagnetism in a metallic object (e.g., object 110c) may result in a pronounced skin effect displacing the induced eddycurrents into a thin layer (skin) at the surface of the object. This maysubstantially reduce the effective electrical conductivity of the objectcausing substantially higher power dissipation if compared to anon-ferromagnetic metallic object. Further, lengthy ferromagnetic,metallic objects (e.g., objects 110 d) that may reflect an impedanceΔZ_(r) in the angle range 608 tend to experience magnetic saturationresulting in excessive hysteresis losses and consequent heating.Therefore, this object category may be characterized by the highest losspower density (e.g., in Watt per unit surface area) and thus highestheating temperatures. Therefore, it may be desirable to selectivelyincrease a sensitivity of a multi-purpose detection circuit 100 toobjects (e.g., objects 110) of this category as disclosed in U.S. Pat.No. 10,495,773 titled Improving Foreign Object Detection forFerromagnetic Wire-Like Objects, the entire contents of which are herebyincorporated by reference.

In another aspect of the multi-purpose detection circuit 100, theinductive sense circuit (e.g., inductive sense circuit 501 of FIG. 5Ausing sense coil 502) may be used for capacitive sensing of livingobjects (e.g., a human hand, a cat, or any other animal) that arepredominantly dielectric and that may be located in proximity of thesense coil. Such use case may require the multi-purpose detectioncircuit 100 to be able to discriminate dielectric objects (e.g., object112 or 114) from metallic objects (e.g., object 110). Suchdiscrimination may be required, if measures upon detection of adielectric object (e.g., object 112 or 114) differ from those appliedupon detection of a metallic object.

FIGS. 7A to 7I illustrate example implementations of another portion ofthe multi-purpose detection circuit 100 of FIG. 1 based on capacitivesensing by measuring at least one electrical characteristic (e.g., acomplex impedance). These examples are to illustrate the principle ofthe sensing and measurement technique and do not show all the details ofa multi-purpose detection circuit 100. Particularly, for illustrativepurposes, they only show a single capacitive sense circuit rather thanthe plurality of capacitive sense circuits (e.g., the plurality ofcapacitive sense circuits 108 a, 108 b, . . . , 108 n with reference toFIG. 1). Further, they do not show the details of the signal generation,signal processing, and evaluation as it may be required e.g., fordetermining at least one of a presence of a foreign object, a livingobject, a vehicle, a type of vehicle, and a position of the vehicle andas illustrated by the block diagram of FIG. 4.

The descriptions of the circuits 700, 710, 720, 730, 740, and 750 ofFIGS. 7A to 7F, respectively, are based on measuring a one-portimpedance Z₁₁, while the circuits 760, 770, and 780 of FIGS. 7F to 7I,respectively, employ a two-port transimpedance Z₂₁ measurement at thesense frequency e.g., using a sinusoidal sense signal. However, thisshould not exclude implementations configured to measure otherelectrical characteristics using other sense signal waveforms (e.g.,multi frequency signals, pulse signals, pseudo random signals, etc.).

In some implementations, the sense signal is a high frequency signalwith a spectrum substantially in the MHz range (e.g., in a range from2.5 MHz to 3.5 MHz). In other implementations, the sense signal isconstraint to the range from 3.155 MHz to 3.400 MHz for frequencyregulatory reasons as previously mentioned in connection with FIGS. 5Ato 5F. In some geographic regions or countries, this frequency range maypermit higher magnetic field strength level H in the specified distancefrom the plurality of capacitive sense elements (e.g., the plurality ofcapacitive sense elements 109 of the multi-purpose detection circuit100).

The ground symbol shown in the schematic diagrams of FIGS. 7A to 7Iindicate a network node on ground potential referred to as the “circuitground”. However, this should not exclude non-ground-basedimplementations or implementations that use different grounds ondifferent potentials.

The circuit 700 of FIG. 7A illustrates an example implementation basedon measuring a complex impedance Z₁₁ of a one-port capacitive sensecircuit 701 (shown in FIG. 7A as the circuit on the right side of thedashed line). More specifically, the impedance Z₁₁ is measured at themeasurement port 708 (indicated in FIG. 7A by a terminal and a dashedline) by applying, from the current source 512, a sinusoidal current I₀at the sense frequency (e.g., in the MHz range) with a defined amplitudeand phase and by measuring, using a voltage measurement circuit 510, thecomplex open-circuit voltage V (amplitude and phase) as previouslydescribed with reference to FIG. 5A.

The sense circuit 701 comprises a single-electrode capacitive senseelement 702 (single-ended sense electrode 702) having a signal terminal703, a capacitance C and an equivalent series resistance R, a seriesinductor 704 having an inductance L_(s) and an equivalent seriesresistance R_(Ls) electrically connected in series to the senseelectrode 702 at the signal terminal 703, and a parallel inductor 706having an inductance L_(p) and an equivalent series resistance R_(Lp)electrically connected to the series inductor 704 and in parallel to themeasurement port 708.

It may be appreciated that electrical losses in the series inductor 704and in the parallel inductor 706 are the most prominent losses in thecapacitive sense circuit 701. These losses may prevail the electricallosses intrinsic to the sense electrode 702 and extraneous losses in itssurrounding materials (e.g., the Litz wire of the WPT coil 202, theferrite, and the plastic housing of the wireless power transferstructure 200 where the sense electrode 702 may be integrated). Thesematerials may interact with the predominantly electric field asgenerated by the sense electrode 702 causing some losses that may beincluded in the equivalent series resistance R as indicated in FIG. 7A.

The sense electrode's 702 capacitance C may include various capacitancesas indicated in FIG. 7A by dashed lines. Particularly, it may includecapacitance C_(eg) of the sense electrode 702 towards ground and acapacitance C_(ew) towards the WPT coil 202 with reference to FIG. 2.The circuit 700 further illustrates the sense signal current source 512and the voltage measurement circuit 510 both electrically connected tothe sense circuit 701 at the measurement port 708.

The sense electrode 702 may also include a self-inductance (notindicated in FIG. 7A). The associated magnetic fields may interact witha metallic object (e.g., object 110). However, this effect may beinsignificant compared to that of the electric field that also interactswith a metallic object (e.g., object 110).

The sense circuit 701 may be configured to provide a local minimum inthe impedance magnitude |Z_(11,0)(ω)| (series resonance) substantiallyat the nominal sense frequency (e.g., at 3 MHz), where Z_(11,0) refersto the impedance as presented by the sense circuit 701 at themeasurement port 708 in absence of a foreign object with reference toFIG. 3. Alternatively, the sense circuit 701 may be configured toprovide a local minimum of the admittance magnitude function|Y_(11,0)(ω)| (parallel resonance) substantially at the nominal sensefrequency, where Y_(11,0) (=1/Z_(11,0)) refers to the admittance aspresented by the sense circuit 701 at the measurement port 508 inabsence of a foreign object.

In an example series resonant configuration of the sense circuit 701,the reactance of the series inductor 704 substantially compensates forthe reactance of the sense electrode 702 at the nominal sense frequencyproviding an impedance Z_(11,0) that is substantially real (resistive).In this configuration, the inductance L_(p) of the parallel inductor 706may be similar or larger than the inductance L_(s) of the seriesinductor 704. In other terms, the impedance magnitude of the parallelinductor 706 may be substantially (e.g., 10 times) higher than theimpedance magnitude |Z_(11,0)| as presented at the nominal sensefrequency. In this configuration, the parallel inductor 706 may exert anegligible impact on the impedance |Z_(11,0)| at the nominal sensefrequency.

In an example parallel resonant configuration of the sense circuit 701,the reactance of the series inductor 704 undercompensates for thereactance of the sense electrode 702 at the nominal sense frequency. Theresidual capacitive susceptance of the series connection of the inductor704 and the sense electrode 702 is substantially compensated for by thesusceptance of the parallel inductor 706 providing an admittanceY_(11,0) that is substantially real (resistive). In this configuration,the inductance L_(p) of the parallel inductor 706 may be smaller,similar, or larger than the inductance L_(s) of the series inductor 704.Stated in other terms, the admittance magnitude of the parallel inductor706 may be substantially (e.g., 20 times) higher than the admittancemagnitude |Y_(11,0)| as presented at the nominal sense frequency. Inthis configuration, the parallel inductor 706 exerts a significantimpact on the admittance Y_(11,0) at the nominal sense frequency.

In some implementations, the parallel inductor 706 together with theseries inductor 704 are used for purposes of resonance tuning andimpedance transformation, e.g., to transform the impedance Z₁₁ to matchthe sense circuit 701 with an operating impedance range as previouslymentioned with reference to FIG. 1. The inductance ratio L_(s)/L_(p) maybe a parameter to control the impedance magnitude |Z_(11,0)|.

Impedance transformation may be particularly effective, if the sensecircuit 701 is configured for parallel resonance. More specifically,increasing the inductance ratio L_(s)/L_(p), while maintaining parallelresonance at the nominal sense frequency, may substantially increase theadmittance |Y_(11,0)| of the parallel resonant configuration at thenominal sense frequency.

Increasing the inductance ratio L_(s)/L_(p), while maintaining resonanceat the nominal sense frequency, may also somewhat decrease the impedance|Z_(11,0)| as presented at the nominal sense frequency in the seriesresonant configuration of the sense circuit 701. However, impedancetransformation may be limited and far less effective than that of theparallel resonant configuration.

In another aspect of resonance tuning, at least one of the seriesinductor 704 and the parallel inductor 706 include a variable inductoras previously discussed with reference to FIG. 5A. In someimplementations of the circuit 70, at least one of the variableinductors 704 and 706 is used to compensate for a temperature drift, anageing, or a detuning of the sense circuit 701 caused by an externalimpact and to maintain its resonance substantially at the nominal sensefrequency. In a further aspect, the variable inductor 704 in combinationwith the variable inductor 706 are used to vary the impedance |Z_(11,0)|of the sense circuit 711.

In yet another aspect, the sense electrode's 702 capacitance C incombination with the parallel inductor 706 form a 2^(nd) order high passfilter to attenuate a low frequency disturbance component in the voltageV for purposes as previously discussed with reference to FIG. 5A. Thislow frequency disturbance component may emanate from a disturbancecurrent capacitively coupled into the sense electrode 702 (e.g., viacapacitance C_(ew)) during wireless power transfer.

With reference to FIG. 1, the sense circuit 701, the sense electrode702, and the series inductor 704 may correspond e.g., to the capacitivesense circuit 108 a, the capacitive sense element 109 a (comprising adouble-ended sense electrode that may be electrically connected inparallel to form a single-ended sense electrode), and the associatedinductive element, respectively. The current source 512 may include thesignal generator circuit 406 and the driver circuit 402, while thevoltage measurement circuit 510 may include the measurement amplifiercircuit 404 and the signal processing circuit 408 with reference to FIG.4.

In some implementations, the current source 512 may be characterized bya quasi-ideal current source and the voltage measurement circuit 510 bya quasi-ideal voltage measurement circuit as previously defined withreference to FIG. 5A.

Though not shown herein, other impedance measurement techniques (e.g.,the voltage source current measurement technique) may also becontemplated as previously discussed with reference to the circuit 500of FIG. 5A.

Further, in some implementations, measurement of the voltage V and thusof the impedance Z₁₁ may be affected by noise and other disturbancesignals reducing a detection sensitivity of the multi-purpose detectioncircuit 100. The noise may include circuit intrinsic noise as generatedin active and passive components of the circuit 700 of FIG. 7A. It mayalso include quantization noise e.g., generated in a digitalimplementation of the signal generator circuit 406 and the signalprocessing circuit 408 with reference to FIG. 4. Other disturbancesignals may emanate from sources external to the circuit 700 (e.g., fromthe WPT system during wireless power transfer, from a switched-modepower supply, from a digital processing unit, etc.). These circuitextrinsic disturbance signals may be capacitively coupled (e.g., viacapacitance C_(ew)) to the sense electrode 702 and may include thefundamental and harmonics of the WPT operating frequency and otherswitching noise components as generated by the WPT system. Therefore, insome implementations, the voltage measurement circuit 510 includes afilter to selectively filter the sense signal and to suppress noise andother disturbance signal components as previously discussed withreference to FIG. 5A

Moreover, in implementations employing a selective voltage measurementcircuit 510 as discussed above, the sense signal waveform as generatedby the current source 512 and the corresponding filter of the voltagemeasurement circuit 510 are adapted e.g., to improve the SNR andconsequently to improve the detection sensitivity as previouslydiscussed with reference to FIG. 5A.

With reference to FIG. 1, FIG. 7A also illustrates the non-livingobjects 110 and 112 and the living object 114 proximate to the senseelectrode 702. Presence of the object 110, 112, 114, or vehicle 330 maycause a change in one or more electrical characteristics of the sensecircuit 701. As non-limiting examples, it may cause a change in thecapacitance C and in the equivalent series resistance R resulting in animpedance change ΔZ with respect to the impedance Z_(11,0) as measuredin absence of a foreign object with reference to FIG. 3. Presence of anobject (e.g., object 114) may be determined if ΔZ satisfies certaincriteria (e.g., the magnitude of ΔZ exceeds a detection threshold).Though not shown in FIG. 7A, a change ΔZ in the measured impedance Z₁₁may also be caused by the underbody of a vehicle or by the vehicle-basedwireless power transfer structure (e.g., vehicle 330 and vehicle-basedwireless power transfer structure 310 with reference to FIG. 3), whichmay indicate presence of a vehicle above the sense electrode 702.Further, an impedance change ΔZ may also be caused by a substantiallyconductive (metallic) object (e.g., object 110) proximate to the senseelectrode 702 since it also interacts with the electric field asgenerated by the sense electrode 702. Stated in other terms, a metalobject (e.g., object 110) proximate to the sense electrode 702 maychange one or more the capacitances C_(eg) and C_(eW) as illustrated inFIG. 7A as well as the self-inductance as previously mentioned.

In an implementation of the circuit 700 based on measuring theadmittance Y₁₁, presence of the object 110, 112, 114, or vehicle 330 maycause a change ΔY with respect to the admittance Y_(11,0) as measured inabsence of a foreign object. Analogously, presence of an object (e.g.,object 110) may be determined if ΔY satisfies certain criteria (e.g.,the magnitude of ΔY exceeds a detection threshold).

As previously discussed with reference to the circuit 500 of FIG. 5Ausing a quasi-ideal current source 512, a change ΔZ in the impedance Z₁₁(e.g., due to the presence of the object 114) manifests in a change ΔVin the voltage V that is proportional to ΔZ while the current I₀ remainssubstantially unaffected. Therefore, measuring the complex voltage V maybe equivalent to measuring the complex impedance Z₁₁ and there may be norequirement for additionally measuring the current I thus reducingcomplexity of the measurement circuit (e.g., measurement circuit 104 ofFIG. 1)

With reference to Equation (8) and (9), the fractional change ΔZ′ (orΔY′) caused by a defined test object (e.g., object 112) placed at adefined position relative to the sense electrode 702 may relate to thedetection sensitivity of an object detection circuit (e.g., themulti-purpose object detection circuit 100 of FIG. 1) based on theone-port capacitive sense circuit 701. More specifically, increasing thefractional change ΔZ′ (or ΔY′) may increase the SNR as defined byEquation (10). As non-limiting examples, the fractional change ΔZ′ (orΔY′) may be increased by optimizing the design of the sense electrode702 with respect to its geometry and its integration into the wirelesspower transfer structure (e.g., wireless power transfer structure 200with reference to FIGS. 2 and 3), by resonance tuning e.g., using theseries inductor 704, and by improving the Q-factor of the sense circuit701. Improving the Q-factor may increase the SNR, if the noise voltageV_(n) is predominantly circuit intrinsic noise as discussed below withreference to FIG. 7J.

As further analyzed and discussed below with reference to FIG. 7J, useof the parallel inductor 506 for purposes of parallel resonance tuningand impedance transformation may result in a lower fractional change ifcompared to the fractional change of the series resonant configurationof the sense circuit 701.

As previously discussed with reference to the circuit 500 of FIG. 5A, itmay be desirable to discriminate between certain categories of objectse.g., between living objects (e.g., object 114) and non-living objects(e.g., object 112). In another aspect, it may also be desirable todiscriminate e.g., between living objects (e.g., object 114) and thevehicle 330 with reference to FIG. 3. As further discussed below withreference to FIG. 8A, this may be accomplished based on characteristicsof the change of the sense electrode's 702 impedance as produced by anyof the objects 110, 112, 114, or vehicle 330, also referred to herein asthe reflected impedance ΔZ_(r). As further discussed below withreference to FIG. 8A, the reflected impedance ΔZ_(r) and particularlythe angle arg{ΔZ_(r)} may reflect electrical properties of the object110, 112, 114, or vehicle 330. The same is true for the reflectedadmittance ΔY_(r).

In some implementations and configurations of the circuit 700 of FIG.7A, the change ΔZ in the impedance Z₁₁ caused by an object (e.g., object114) is indicative of the reflected impedance ΔZ_(r). Therefore, in anaspect of object discrimination, the circuit 700 may be configured todetermine the angle arg{ΔZ} with the required accuracy. However, in someimplementations, measuring the angle arg{ΔZ} may be subject to errorsfor various reasons as previously discussed with reference to FIG. 5A.

In an aspect of reducing an error in the measurement of the anglearg{ΔZ}, some implementations of a multipurpose detection circuit 100employ a phase calibration of the analog circuitry (e.g., the analogfront end portion of the measurement circuit 104 with reference to FIG.4) as previously described with reference to FIG. 5A.

Reactance compensation (resonance tuning) in the sense circuit 701produces a local extremum (minimum or maximum) in the impedancemagnitude function |Z_(11,0)(ω)| and hence in the voltage magnitude |V|across the measurement port 708. Therefore, reactance compensationprovides a mean to calibrate the voltage measurement circuit 510 andhence the impedance measurement with respect to the angle arg{Δz}.

In a first step of an example calibration procedure applicable to theseries resonant configuration of the circuit 700 of FIG. 7A, the sensefrequency is adjusted to the local minimum of the voltage magnitude |V|as measured by the voltage measurement circuit 510 supposing absence ofa foreign object. At this frequency, the complex impedance Z_(11,0) andhence the complex voltage V across the measurement port 708 may besubstantially real. Otherwise stated, the angles arg{Z_(11,0)} andarg{V} are substantially zero. In a second step of the examplecalibration procedure, the voltage measurement circuit 510 is correctedby applying a phase shift (impedance plane rotation) as previouslydescribed with reference to FIG. 5A and defined by Equation (13).

Applying the angle correction of Equation (13), an object (e.g., object114) reflecting an impedance ΔZ_(r) that is imaginary (reactive) maycause a measured voltage change ΔV_(cal) that is substantiallyimaginary. Nevertheless, a small residual error may remain in the anglearg{ΔV_(cal)} due to the impact of the parallel inductor 706 and theelectrical losses in the sense circuit 701. The residual angle error ofan example series resonant configuration of the circuit 700 and for anexample object 114 is provided in TABLE 4.

In some implementations, the residual error described above is reducedby configuring the parallel inductor 706 with an inductance L_(p) whoseimpedance Z_(Lp) is substantially larger (e.g., 10 times larger) thanthe series resonant resistance of the sense circuit 701. In otherimplementations, the residual error is reduced by measuring theimpedance Z_(11,0) at two or more substantially different frequenciesand by determining the elements of an equivalent circuit model of thesense circuit 701 (e.g., the equivalent circuit model illustrated inFIG. 7J) based on the measured impedances Z_(11,0) employing a best fitmethod. In some implementations, these two or more frequencies includeat least the frequency of the minimum and the maximum of |Z_(11,0)(ω)|.

In an implementation of the multipurpose detection circuit 100 using aplurality of capacitive sense circuits (e.g., capacitive sense circuits108 a, 108 b, . . . , 108 n), each including a respective capacitivesense element (e.g., capacitive sense element 109 a, 109 b, . . . , 109n), a residual error may be caused by a parasitic resonance effect ofsense circuits associated to adjacent capacitive sense elements of anarrangement of sense electrodes. More precisely, a residual error in afirst sense circuit (e.g., capacitive sense circuit 108 a) including afirst capacitive sense element (e.g., capacitive sense element 109 a)may be caused by a parasitic resonance effect of at least one secondcapacitive sense circuit (e.g., capacitive sense circuit 108 b)including a second capacitive sense element (e.g., capacitive senseelement 109 b) that is located adjacent to the first capacitive senseelement.

Therefore, in some implementations of the multipurpose detection circuit100, the measurement accuracy of the angle arg{ΔZ} and thus of the anglearg{ΔZ_(r)} is increased by an optimized design of the sense electrode702 and by introducing some spacing between adjacent sense electrodes702 in an arrangement of sense electrodes.

In an implementation configured for parallel resonance as defined above,the circuit 700 may be configured to measure the admittance Y₁₁ andcorresponding changes ΔY of Y₁₁ as caused by the object 110, 112, 114,or vehicle 330. In this case, the admittance change ΔY may be indicativeof the reflected impedance ΔZ_(r) as previously introduced. As discussedabove with reference to the series resonant configuration, the anglearg{ΔY} may be subjected to an error and therefore may requirecalibration to reduce an error in the measurement of the angle arg{ΔY}and thus of the angle arg{ΔZ_(r)}.

In an implementation configured for parallel resonance, the circuit 700may be calibrated analogously to the series resonant configurationhowever using the local minimum of the admittance function |Y_(11,0)(ω)|where susceptance compensation occurs.

In a first step of an example calibration procedure applicable to theparallel resonant configuration of the circuit 700 of FIG. 7A, the sensefrequency is adjusted to the local maximum of the voltage magnitude |V|as measured by the uncalibrated voltage measurement circuit 510supposing absence of a foreign object. At this frequency, the admittanceY_(11,0) and hence the voltage V across the measurement port 708 may besubstantially real. Otherwise stated, the angles arg{Y_(11,0)} andarg{V} are substantially zero. In a second step of the examplecalibration procedure, the voltage measurement circuit 510 is correctedby applying a phase shift (impedance plane rotation) as defined above byEquation (13).

Applying the angle correction of Equation (13), an object (e.g., object114) reflecting an impedance ΔZ_(r) that is imaginary (reactive)produces a measured voltage change ΔV_(cal) that is substantiallyimaginary. A residual error may remain in the angle arg{ΔV_(cal)} due tothe transformation of ΔZ_(r) to ΔY in the lossy sense circuit (e.g.,sense circuit 701). The residual angle error of an example parallelresonant configuration of the circuit 700 and for an example reflectedimpedance ΔZ_(r) is provided in TABLE 4.

In an example implementation, the residual error due to thetransformation of ΔZ_(r) to ΔY is reduced by measuring the admittanceY_(11,0) at two or more substantially different frequencies, supposingabsence of a foreign object, and by determining the elements of anequivalent circuit model (e.g., the equivalent circuit model of FIG. 7J)based on the measured admittances Y_(11,0) employing a best fit method.In some implementations, these two or more frequencies include at leastthe frequency of the minimum and the maximum of |Y_(11,0)(ω)|.

The series and the parallel resonant configuration of the circuit 700 ofFIG. 7A are analyzed below with reference to FIG. 7J with respect tovarious characteristics such as the Q-factor, fractional change, andvarious definitions of SNR based on an equivalent circuit model.

The circuit 710 of FIG. 7B illustrates another example implementationbased on measuring a complex impedance Z₁₁ of a one-port capacitivesense circuit 711 (shown in FIG. 7B as the circuit on the right side ofthe dashed line). More specifically, the impedance Z₁₁is measured at themeasurement port 718 (indicated in FIG. 7B by a terminal and a dashedline) by applying, from the current source 512, a sinusoidal current I₀and by measuring, using the voltage measurement circuit 510, the complexopen-circuit voltage V as previously described with reference to FIG.5A.

As the sense circuit 701 of FIG. 7A, the sense circuit 711 comprises thesingle-electrode capacitive sense element 702 (single-ended senseelectrode 702) having the signal terminal 703 and the capacitance C withreference to FIG. 7A, a series inductor 714 having an inductance L_(s)electrically connected in series to the sense electrode 702, and aparallel inductor 716 having an inductance L_(p) electrically connectedto the series inductor 714 and in parallel to the measurement port 718.The capacitance C may include the capacitances C_(eg) and C_(ew) (notshown in FIG. 7B) as previously discussed with reference to FIG. 7A. Thesense circuit 711 further includes a parallel capacitor 715 having acapacitance C_(p). The circuit 710 further illustrates the sense signalcurrent source 512 and the voltage measurement circuit 510 electricallyconnected to the measurement port 718.

Though not shown in FIG. 7B for purposes of illustration, the inductiveand capacitive elements of the sense circuit 711 may also causeelectrical losses that may be represented by a respective equivalentseries resistance.

As with the sense circuit 701 of FIG. 7A, the sense circuit 711 may beconfigured to provide a local minimum in the impedance magnitudefunction |Z_(11,0)(ω)| (series resonance) substantially at the nominalsense frequency. Alternatively, the sense circuit 711 may be configuredto provide a local minimum of the admittance magnitude function|Y_(11,0)(ω)| (parallel resonance) substantially at the nominal sensefrequency.

In an example series resonant configuration of the sense circuit 711,the reactance of the series inductor 714 substantially compensates forthe reactance of the sense electrode 702 in parallel to the capacitor715 at the nominal sense frequency providing an impedance Z_(11,0) thatis substantially real (resistive). In this configuration, the inductanceL_(p) of the parallel inductor 706 may be similar or larger than theinductance L_(s) of the series inductor 714. In other terms, theimpedance magnitude of the parallel inductor 716 may be substantially(e.g., 10 times) higher than the impedance magnitude |Z_(11,0)| aspresented at the nominal sense frequency. In this configuration, theparallel inductor 716 may exert a negligible impact on the impedance|Z_(11,0)| at the nominal sense frequency.

In an example parallel resonant configuration of the sense circuit 711,the reactance of the series inductor 714 undercompensates for thereactance of the sense electrode 702 in parallel to the capacitor 715 atthe nominal sense frequency. The residual capacitive susceptance of theseries connection of the inductor 704 and the parallel connection of thesense electrode 702 and capacitor 715 is substantially compensated forby the susceptance of the parallel inductor 716 providing an admittanceY_(11,0) that is substantially real (resistive). In this configuration,the inductance L_(p) of the parallel inductor 706 may be smaller,similar, or larger than the inductance L_(s) of the series inductor 714.Stated in other terms, the admittance magnitude of the parallel inductor716 may be substantially (e.g., 20 times) higher than the admittancemagnitude |Y_(11,0)| as presented at the nominal sense frequency. Inthis configuration, the parallel inductor 716 exerts a significantimpact on the admittance Y_(11,0) at the nominal sense frequency.

In some implementations, the parallel inductor 716 together with theseries inductor 714 and the parallel capacitor 715 are used for purposesof resonance tuning and impedance transformation, e.g., to transform theimpedance Z₁₁ to match the sense circuit 711 with an operating impedancerange as previously mentioned with reference to FIG. 1. The inductanceratio L_(s)/L_(p) and the capacitance ratio C/C_(p) may be parameters tocontrol the impedance magnitude |Z_(11,0|.)

If configured for series resonance, the impedance magnitude |Z_(11,0)|may be decreased mainly by decreasing the capacitance ratio C/C_(p). Ifconfigured for parallel resonance, the admittance magnitude |Y_(11,0)|may be increased mainly by increasing the inductance ratio L_(s)/L_(p).

In another aspect of resonance tuning, the parallel capacitor 715 mayinclude a variable capacitor whose capacitance C_(s) can beelectronically controlled (e.g., a DC controlled capacitor) forming avariable capacitor 715. In some implementations of the circuit 700, avariable capacitor 715 is used to compensate for a temperature drift, anageing, or a detuning of the sense circuit 701 caused by an externalimpact and to maintain its resonance substantially at the nominal sensefrequency. In a further aspect, the variable capacitor 714 incombination with a variable inductor 714 are used to vary the impedance|Z_(11,0)| of the sense circuit 701.

In a further aspect, the sense electrode's 702 capacitance C incombination with the parallel inductor 716 form a 2^(nd) order high passfilter to attenuate a low frequency disturbance component in the voltageV for purposes as previously discussed with reference to FIGS. 5A.

With reference to FIG. 1, FIG. 7B also illustrates the objects 110, 112,and 114 proximate to the capacitive sense element 702. As previouslydiscussed with reference to FIG. 7A, presence of the object 110, 112,114, or vehicle 330 may cause a change in one or more electricalcharacteristics of the sense circuit 711.

The circuit 720 of FIG. 7C illustrates a further example implementationbased on measuring a complex impedance Z₁₁ of a one-port capacitivesense circuit 721 (shown in FIG. 7C as the circuit on the right side ofthe dashed line). More specifically, the impedance Z₁₁ is measured atthe measurement port 728 (indicated in FIG. 7C by a terminal and adashed line) by applying, from the current source 512, a sinusoidalcurrent I₀ and by measuring, using the voltage measurement circuit 510,the complex open-circuit voltage V as previously described withreference to FIG. 7A.

As the sense circuit 701 of FIG. 7A, the sense circuit 721 comprises thesingle-electrode capacitive sense element 702 (also referred to hereinas a single-ended sense electrode) having the capacitance C withreference to FIG. 7A, a series inductor 724 having an inductance L_(s)electrically connected in series to the single-ended sense electrode702. However, the sense circuit 721 shows the parallel inductor 706replaced by a transformer 726 with a transformation ratio n_(T):1 asindicated in FIG. 7C. As previously described with reference to FIG. 5B,the transformer may include a primary winding and a galvanicallyinsulated secondary winding wound on a common core as suggested by thetransformer symbol in FIG. 7C. However, other transformerimplementations may apply as previously mentioned with reference to FIG.5B. FIG. 7C also indicates the main inductance L_(m), the series(leakage) inductance L_(σ), and the equivalent series resistances R_(Lm)and R_(W) referring to the equivalent circuit model of a non-idealtransformer illustrated in FIG. 5H. FIG. 7C shows its primary windingelectrically connected in parallel to the measurement port 728, whileits secondary winding is electrically connected to the series inductor724. The circuit 720 further illustrates the sense signal current source512 and the voltage measurement circuit 510 both electrically connectedto the sense circuit 721 at the measurement port 728.

Though not indicated in FIG. 7C for purposes of illustration, the seriesinductor 724 and the sense electrode 702 may also cause electricallosses that may be represented by a respective equivalent resistance aspreviously discussed with reference to FIG. 7A.

The sense circuit 721 may be configured to provide a local minimum inthe impedance magnitude function |Z_(11,0)(ω)| (series resonance)substantially at the nominal sense frequency. Alternatively, it may beconfigured to provide a local minimum in the admittance magnitudefunction |Y_(11,0)(ω)| (parallel resonance) substantially at the nominalsense frequency using the transformer's 726 secondary referred maininductance L_(m) in a manner similar to using the inductance L_(p) asdescribed above with reference to FIG. 7A.

In an example series resonant configuration of the sense circuit 721,the reactance of the series inductor 724 together with the transformer's726 secondary referred leakage inductance L_(σ) substantiallycompensates for the reactance of the sense electrode 702 at the nominalsense frequency providing an impedance Z_(11,0) at the measurement port728 that is substantially real (resistive). In this configuration, thetransformer's 726 secondary referred main inductance L_(m) may besimilar or larger than the inductance L_(s) of the series inductor 724.Stated in other terms, the primary referred open-circuit impedance ofthe transformer 726 may be substantially (e.g., 10 times) higher thanthe impedance magnitude |Z_(11,0)| as presented at the nominal sensefrequency. Apart from the impedance transformation by the factor n_(T)², the transformer 726 may exert a negligible impact on the impedance|Z_(11,0)| at the nominal sense frequency.

In an example parallel resonant configuration of the sense circuit 721,the reactance of the series inductor 724 together with the transformer's726 secondary referred leakage inductance L_(σ) undercompensates for thereactance of the sense electrode 702 at the nominal sense frequency. Theresidual capacitive susceptance of the series connection of the inductor724, the transformer's 726 leakage inductance L_(σ), and the senseelectrode 702 is substantially compensated for by the susceptance of thetransformer's 726 secondary referred inductance L_(m) providing anadmittance Y_(11,0) that is substantially real (resistive). In thisconfiguration, the inductance L_(m) may be smaller, similar, or largerthan the inductance L_(s) of the series inductor 724. Stated in otherterms, the primary referred open-circuit admittance of the transformer726 may be substantially (e.g., 20 times) higher than the admittancemagnitude |Y_(11,0)| as presented at the nominal sense frequency. Inthis configuration and apart from the admittance transformation, thetransformer 726 exerts a significant impact on the admittance Y_(11,0)at the nominal sense frequency.

The transformer 726 may serve for various purposes. In someimplementations, the transformer 726 is a n_(T):1 transformer with n_(T)#1 used at least for impedance transformation e.g., to match theimpedance magnitude |Z₁₁| of the sense circuit 721 with an operatingimpedance range as previously mentioned with reference to FIG. 5A. In anexample implementation configured for series resonance, the transformer726 increases the impedance |Z₁₁| by a factor n_(T) ² with n_(T)>1. Inanother example implementation configured for parallel resonance, itincreases the admittance |Y₁₁| by a factor 1/n² with n <1. In yet otherimplementations, it is a balancing transformer used to reduce a leakagecurrent e.g., on the feeder cable of the wireless power transferstructure (e.g., wireless power transfer structure 200 of FIGS. 2 and 3)where the sense electrode 702 is integrated. Reducing this leakagecurrent may reduce an unwanted sensitivity of other WPT system parts toa living object (e.g., living object 114). In yet anotherimplementation, the transformer 726 is also part of the resonance tuningusing its main inductance L_(p) in a manner similar to the parallelinductor 506 with reference to FIG. 5A.

Apart from the transformation ratio n_(T):1, the inductance ratioL_(s)/L_(m) may be an additional parameter to match the admittancemagnitude |Y₁₁| of the parallel resonant configuration with an operatingadmittance range of the multi-purpose object detection circuit 100 withreference to FIG. 1.

In a further aspect, the sense electrode's 702 capacitance C incombination with the transformer's 726 main inductance L_(m) form a2^(nd) order high pass filter to attenuate a low frequency disturbancecomponent in the voltage V for purposes as previously discussed inconnection with FIG. 5A.

With reference to FIG. 1, FIG. 7C also illustrates the objects 110, 112,and 114 proximate to the sense electrode 702. As previously discussedwith reference to FIG. 7A, presence of the object 110, 112, 114, orvehicle 330 may cause a change in one or more electrical characteristicsof the sense circuit 741 as previously discussed with reference to FIG.5A.

The circuit 730 of FIG. 7D illustrates yet another exampleimplementation based on measuring a complex impedance Z₁₁ of a one-portcapacitive sense circuit 731 (shown in FIG. 7D as the circuit on theright side of the dashed line). More specifically, the impedance Z₁₁ ismeasured at the measurement port 738 (indicated in FIG. 7D by a terminaland a dashed line) by applying, from the current source 512, asinusoidal current I₀ and by measuring, using the voltage measurementcircuit 510, the complex open-circuit voltage V as previously describedwith reference to FIG. 5A.

As opposed to the sense circuits 701, 711, and 721, the sense circuit731 is operated in a differential mode and uses a substantiallysymmetric double-electrode capacitive sense element (e.g., double-endedsense electrode 732) composed of electrodes 732 a and 732 b (alsoreferred to herein as a double-ended sense electrode) providing adifferential-mode capacitance C. The sense circuit 731 may be split intoa first branch and a second branch with an equal topology. The sensecircuit 731 may be substantially symmetric (balanced) with respect toits capacitances and inductances. Further, the sense circuit 731comprises a differential-mode series inductor 734 having an inductanceL_(s)/2 in each branch and that is electrically connected to thedouble-ended sense electrode 732. Moreover, the sense circuit 731comprises a transformer 736 with a transformation ratio n_(T):1 andsecondary referred main inductance L_(m) with reference to FIG. 5H. Itsprimary winding is electrically connected in parallel to the measurementport 738, while its secondary winding is electrically connected to thedifferential-mode series inductor 734. The circuit 730 furtherillustrates the sense signal current source 512 and the voltagemeasurement circuit 510 both electrically connected to the sense circuit731 at the measurement port 738.

The double-ended sense electrode 732 provides a differential-modecapacitance C that may include various capacitances as indicated in FIG.7D by dashed lines. In particular, it may include an interelectrodecapacitance C_(ee) between electrodes 732 a and 732 b, a capacitanceC_(eg,a) and C_(eg,b) towards ground and a capacitance C_(ew,a) andC_(ew,b) towards the WPT coil 202 for the respective electrode 732 a and732 b.

Though not indicated in FIG. 7D for purposes of illustration, thecapacitive and inductive elements of the sense circuit 731 may causeelectrical losses that may be represented by a respective equivalentseries resistance as previously discussed with reference to FIG. 7A.

The sense circuit 731 may be configured to provide a local minimum inthe impedance magnitude function |Z_(11,0)(ω)| (series resonance)substantially at the nominal sense frequency. Alternatively, it may beconfigured to provide a local minimum in the admittance magnitudefunction |Y_(11,0)(ω)| (parallel resonance) substantially at the nominalsense frequency using the transformer's 736 secondary referred maininductance L_(m) as described above with reference to FIG. 7C.

In an example series resonant configuration of the sense circuit 731,the reactance of the differential-mode series inductor 734 together withthe transformer's 736 secondary-referred leakage inductance L_(σ)substantially compensates for the reactance of the double-ended senseelectrode 732 at the nominal sense frequency providing an impedanceZ_(11,0) at the measurement port 738 that is substantially real(resistive). In this configuration, the transformer's 736 secondaryreferred main inductance L_(m) may be similar or larger than theinductance L_(s) of the differential-mode series inductor 734. Stated inother terms, the primary referred open-circuit impedance of thetransformer 736 may be substantially (e.g., 10 times) higher than theimpedance magnitude |Z_(11,0)| as presented at the nominal sensefrequency. Apart from the impedance transformation by the factor n_(T)², the transformer 736 may exert a negligible impact on the impedance|Z_(11,0)| at the nominal sense frequency.

In an example parallel resonant configuration of the sense circuit 731,the reactance of the differential-mode series inductor 734 together withthe transformer's 736 secondary referred leakage inductance L_(σ)undercompensates for the reactance of the double-ended sense electrode732 at the nominal sense frequency. The residual capacitive susceptanceof the series connection of the differential-mode series inductor 734,the transformer's 736 leakage inductance L_(σ), and the double-endedsense electrode 732 is substantially compensated for by the susceptanceof the transformer's 736 secondary referred inductance L_(m) providingan admittance Y_(11,0) that is substantially real (resistive). In thisconfiguration, the inductance L_(m) may be smaller, similar, or largerthan the inductance L_(s) of the differential-mode series inductor 734.Stated in other terms, the primary referred open-circuit admittance ofthe transformer 736 may be substantially (e.g., 20 times) higher thanthe admittance magnitude |Y_(11,0)| as presented at the nominal sensefrequency. In this configuration and apart from the admittancetransformation, the transformer 736 exerts a significant impact on theadmittance Y_(11,0) at the nominal sense frequency.

In a further aspect, the double-ended sense electrode's 732 capacitanceC in combination with the transformer's 736 main inductance L_(m) form a2^(nd) order high pass filter to attenuate a low frequency disturbancecomponent in the voltage V for purposes as previously discussed withreference to FIG. 5A.

In some implementations, the differential-mode series inductor 734 isconfigured to also provide a common mode inductance e.g., to attenuate adisturbance signal component in the voltage V emanating from a commonmode current capacitively and inductively coupled into the double-endedsense electrode 732 (e.g., via capacitances C_(ew,a) and C_(ew,b))during wireless power transfer.

The transformer 736 may serve for various purposes. In someimplementations, the transformer 736 is used to match the impedance Z₁₁of the sense circuit 731 with an operating impedance range as previouslydiscussed with reference to FIG. 5B. In other implementations, it is a1:1 balancing (balun) transformer used to reduce a common modedisturbance current capacitively coupled to the double-ended senseelectrode 732 (e.g., via capacitance C_(ew,a) and C_(ew,b)). In yetfurther implementations, it is a n_(T):1 transformer with n_(T)≠1 andserves for both impedance transformation and balancing.

Apart from the transformation ratio n_(T):1, the inductance ratioL_(s)/L_(m) may be an additional parameter to match the admittancemagnitude |Y_(11,0)| of the parallel resonant configuration with anoperating admittance range of the multi-purpose object detection circuit100 with reference to FIG. 1.

With reference to FIG. 1, the sense circuit 731, the double-ended senseelectrode 732, and the differential-mode series inductor 734 maycorrespond e.g., to the capacitive sense circuit 108 a, the capacitivesense element 109 a (comprising a double-ended sense electrode), and theassociated inductive element, respectively.

With reference to FIG. 1, FIG. 7D also illustrates the objects 110, 112,and 114 proximate to the electrodes 732 a and 732 b. As previouslydiscussed with reference to FIG. 7A, presence of the object 110, 112,114, or vehicle 330 may cause a change in one or more electricalcharacteristics of the sense circuit 731.

In another aspect, the use of a double-ended sense electrode (e.g.,double-ended sense electrode 732) may reduce a disturbance voltagecomponent in the voltage V e.g., emanating from the voltage capacitivelycoupled into the sense electrode by the electric field as generatedduring wireless power transfer. Due to its symmetry, the double-endedsense electrode 732 integrated into a wireless power transfer structure(e.g., wireless power transfer structure 200 of FIG. 1) may pick-upsubstantially less disturbance voltage as compared to an equivalentsingle-ended sense electrode (sense electrode 702 of FIG. 7A) formed byconnecting the electrodes 732 a and 732 b in parallel. On the otherhand, the fractional change defined by Equation (8) and (9) of a sensecircuit (e.g., sense circuit 731) using the double-ended sense electrode732 may be substantially smaller than that of a sense circuit (e.g.,sense circuit 704) using an equivalent single-ended sense electrode 702formed by connecting the electrodes 732 a and 732 b in parallel.Therefore, the resulting SNR (e.g., defined by Equation (10)) asobtained for the sense circuits 731 and 701 when driven with the samecurrent I₀ may not differ that much. However, there may be locations,where the sense circuit 731 using the double-ended sense electrode 732is less sensitive for detecting an object (e.g., living object 114) thanthe sense circuit 701 using an equivalent single-ended sense electrode702.

In a further aspect, the electromagnetic emissions produced by thedouble-ended sense electrode 732 when driving the sense circuit 731 witha current I₀ may be substantially lower than that of the equivalentsingle-ended sense electrode 702 when driving the sense circuit 701 withthe same current I. Therefore, if the drive current to is emissionconstraint (e.g., for frequency regulatory reasons), the sense circuit731 may be driven with a substantially higher current I₀ than the sensecircuit 701 improving the SNR to compete with the sense circuit 701using the single-ended sense electrode 702.

The circuit 740 of FIG. 7E illustrates another example implementationbased on measuring a complex impedance Z₁₁ of a one-port inductive sensecircuit (e.g., sense circuit 741, shown in FIG. 7E as the circuit on theright side of the dashed line). More specifically, the impedance Z₁₁ ismeasured at the measurement port 748 (indicated in FIG. 7E by a terminaland a dashed line) by applying, from a voltage source 552, a sinusoidalvoltage V₀ and by measuring, using a current measurement circuit 550,the complex short-circuit current I as previously mentioned withreference to FIG. 5A (voltage source current measurement technique).

The circuit 740 may be considered as electrically dual to the circuit700 of FIG. 7A. The circuit 740 includes the sense circuit 741comprising the single-ended sense electrode 702 having a capacitance Cand an equivalent series resistance R with reference to FIG. 7A, aparallel inductor 744 having an inductance L_(p) and an equivalentseries resistance R_(Lp), a series capacitor 746 having a capacitanceC_(s) electrically connected in series to the parallel connection of thesense electrode 702 and the parallel inductor 744. The circuit 740further illustrates the sense signal voltage source 552 electricallyconnected to the sense circuit 741 at the measurement port 748 via thecurrent measurement circuit 550. As opposed to FIG. 5C, FIG. 7Eillustrates the current source (e.g., the current measurement circuit550) as non-ground-based (floating).

In an example implementation (not shown herein), the non-ground-basedcurrent source is accomplished by using a ground-based current sourcewith an output transformer providing galvanic isolation.

As previously discussed with reference to the circuit 700 of FIG. 7A,the circuit 740 of FIG. 7E may be configured to be operated at parallelresonance substantially at the nominal sense frequency. Alternatively,it may be configured for series resonance substantially at the nominalsense frequency.

In an example parallel resonant configuration of the sense circuit 741,the susceptance of the parallel inductor 744 substantially compensatesfor the susceptance of the sense electrode 702 at the nominal sensefrequency providing an admittance Y_(11,0) that is substantially real(resistive). In this configuration, the capacitance C_(s) of the seriescapacitor 746 may be similar or larger than the capacitance C of thesense electrode 702. Stated otherwise, the admittance magnitude of theseries capacitor 746 may be substantially (e.g., 10 times) higher thanthe admittance magnitude |Y_(11,0)| as presented at the nominal sensefrequency. In this configuration, the series capacitor 746 may exert anegligible impact on the admittance |Y_(11,0)| at the nominal sensefrequency.

In an example series resonant configuration of the sense circuit 741,the susceptance of the parallel inductor 744 overcompensates for thesusceptance of the sense electrode 702 at the nominal sense frequency.The residual inductive reactance of the parallel connection of theparallel inductor 744 and the sense electrode 702 is substantiallycompensated for by the reactance of the series capacitor 746 providingan impedance Z_(11,0) that is substantially real (resistive). In thisconfiguration, the capacitance C_(s) of the series capacitor 746 may besmaller, similar, or larger than the capacitance C of the senseelectrode 702. Stated otherwise, the impedance magnitude of the seriescapacitor 746 may be substantially (e.g., 20 times) higher than theimpedance magnitude |Z_(11,0)| as presented at the nominal sensefrequency. In this configuration, the series capacitor 746 exerts asignificant impact on the impedance Z_(11,0) at the nominal sensefrequency.

In some implementations, the series capacitor 746 together with theparallel inductor 744 are used for purposes of resonance tuning andimpedance transformation e.g., to transform the impedance Z₁₁ to matchthe sense circuit 741 with an operating impedance range as previouslymentioned with reference to FIG. 1. The capacitance ratio C/C_(s) may bea parameter to control the impedance magnitude |Z_(11,0|.)

Impedance transformation may be particularly effective, if the sensecircuit 741 is configured for series resonance. More specifically,increasing the capacitance ratio C/C_(s), while maintaining seriesresonance at the nominal sense frequency, may substantially increase theimpedance magnitude |Z_(11,0)| at the nominal sense frequency.Therefore, in an aspect, the sense circuit 741 in the series resonantconfiguration may be considered as an alternative to the sense circuit711 of FIG. 7B using the parallel capacitor 715 or to the sense circuit721 of FIG. 7C using the transformer 726.

Increasing the capacitance ratio C/C_(s), while maintaining resonance atthe nominal sense frequency, may also somewhat decrease the admittancemagnitude |Y_(11,0)| as presented at the nominal sense frequency in theparallel resonant configuration of the sense circuit 741. However,impedance transformation may be limited and far less effective than thatof the series resonant configuration.

In a further aspect, the sense electrode's 702 capacitance C incombination with the parallel inductor 744 and the series capacitor 746form a higher order high pass filter for purposes as previouslydiscussed in connection with FIG. 5A.

With reference to FIG. 1, FIG. 7E also illustrates the objects 110, 112,and 114 proximate to the sense electrode 702. As previously discussedwith reference to FIG. 1, presence of the object 110, 112, 114, orvehicle 330 may cause a change in one or more electrical characteristicsof the sense circuit 751. As non-limiting examples, they may cause achange in the capacitance C and in an equivalent series resistance Rthat is considered included in R. This change results in a change ΔZwith respect to the impedance Z_(11,0) as measured in absence of aforeign object with reference to FIG. 3.

The fractional change ΔY′ (or ΔZ′) as defined by Equations (8) and (9)and with respect to a defined test object (e.g., object 112) placed at adefined position relative to the sense electrode 702 may relate to thedetection sensitivity of an object detection circuit (e.g., themulti-purpose detection circuit 100 of FIG. 1) based on the sensecircuit 741. More specifically, increasing the fractional change ΔY′ (orΔZ′) may increase a signal-to-noise ratio (SNR) e.g., as defined byEquation (14).

As non-limiting examples, the fractional change may be increased byoptimizing the design of the sense electrode 702 with respect to itsgeometry and its integration into the wireless power transfer structure(e.g., wireless power transfer structure 200 with reference to FIGS. 2and 3), by resonance tuning e.g., using the parallel inductor 744, andby improving the Q-factor of the sense circuit 741. Improving theQ-factor may increase the SNR, if the noise current I_(n) ispredominantly circuit intrinsic noise as discussed above with referenceto FIG. 5G.

As previously discussed with reference to the circuit 700 of FIG. 7A, itmay be desirable to discriminate between certain categories of objects(e.g., object 110 and 112) e.g., based on the reflected admittanceΔY_(r) that may be indicative of electrical properties of the object110, 112, 114, or vehicle 330.

In some implementations and configurations of the circuit 740 of FIG.7E, the change ΔY in the admittance Y₁₁ caused by an object (e.g.,object 114) is indicative of the reflected admittance ΔY_(r). Therefore,in an aspect of object discrimination, the circuit 740 may be configuredto determine the angle arg{ΔY} and thus the angle arg{ΔY,} with therequired accuracy. However, in some implementations, measuring theadmittance Y₁₁ including the change ΔY may be subject to errors forvarious reasons as previously discussed with reference to the circuit700 of FIG. 7A.

Susceptance compensation in the sense circuit 741 exhibiting a localextremum (minimum or maximum) in the admittance magnitude function|Y_(11,0)(ω)| and hence in the resulting current magnitude |I| at themeasurement port 748 provides a mean to calibrate the currentmeasurement circuit 550 and hence the admittance measurement withrespect to the angle arg{ΔY}.

In a first step of an example calibration procedure applicable to theparallel resonant configuration of the circuit 740 of FIG. 7E, the sensefrequency is adjusted to the local minimum of the current magnitude |I|as measured by the uncalibrated current measurement circuit 550supposing absence of a foreign object. At this frequency, the admittanceY_(11,0) and hence the current I at the measurement port 748 may besubstantially real. Otherwise stated, the angles arg{Y_(11,0)} andarg{I} are substantially zero. In a second step of the examplecalibration procedure, the current measurement circuit 550 is correctedby applying a phase shift such that the imaginary part of the complexcurrent value as determined and output by the current measurementcircuit 550 at this frequency vanishes. Applying the phase shift isequivalent to rotating the admittance plane by an angle arg{I_(uncal)}where I_(uncal) refers to the complex current value as determined by theuncalibrated current measurement circuit 550 (before any correction isapplied). This angle correction may be expressed by Equation (15).

Applying the angle correction of Equation (15), an object (e.g., object114) reflecting an admittance ΔY_(r) that is imaginary (reactive) mayresult in a measured current change ΔI_(cal) that is substantiallyimaginary. Nevertheless, a residual error may remain in the anglearg{ΔI_(cal)} due to the impact of the series capacitor 746 and theelectrical losses in the sense circuit 741. The residual angle error ofan example parallel resonant configuration of the circuit 740 and for anexample object 110 is provided in TABLE 4.

In some implementations, the residual error is reduced by configuringthe series capacitor 746 with a capacitance C_(s) whose admittanceY_(Cs) is substantially larger (e.g., 10 times larger) than the parallelresonant conductance of the sense circuit 741. In other implementations,the residual error is reduced by measuring the admittance Y_(11,0) attwo or more substantially different frequencies and by determining theelements of an equivalent circuit model of the sense circuit 741 (e.g.,the equivalent circuit model illustrated in FIG. 7K) based on themeasured admittances Y_(11,0) employing a best fit method. In someimplementations, these two or more frequencies include at least thefrequency of the minimum and the maximum of |Y_(11,0)(ω)|.

In an implementation configured for series resonance as defined above,the circuit 540 may be configured to measure the impedance Z₁₁ andcorresponding changes ΔZ of Z₁₁ as caused by the object 110, 112, 114,or vehicle 330. In this case, the impedance change ΔZ may be indicativeof the reflected admittance ΔY, as previously introduced. As discussedabove with reference to the parallel resonant configuration, the anglearg{ΔZ} may be subjected to an error and therefore may requirecalibration to reduce an error in the measurement of the angle arg{ΔZ}and thus of the angle arg{ΔY_(r)}.

In an implementation configured for series resonance, the circuit 740may be calibrated analogously to the parallel resonant configurationhowever using the local minimum of the impedance function |Z_(11,0)(ω)|where reactance compensation occurs.

In a first step of an example calibration procedure applicable to theseries resonant configuration of the circuit 740 of FIG. 7E, the sensefrequency is adjusted to the local maximum of the current magnitude |I|as measured by the uncalibrated current measurement circuit 550supposing absence of a foreign object. At this frequency, the impedanceZ_(11,0) and hence the current I at the measurement port 748 may besubstantially real. Otherwise stated, the angles arg{Z_(11,0)} andarg{I} are substantially zero. In a second step of the examplecalibration procedure, the current measurement circuit 550 is correctedby applying a phase shift (impedance plane rotation) as given above byEquation (15).

Applying the angle correction of Equation (15), an object (e.g., object114) reflecting an admittance ΔY_(r) that is imaginary (reactive) mayresult in a measured current change ΔI_(cal) that is substantiallyimaginary. Nevertheless, a residual error may remain in the anglearg{ΔI_(cal)} due to the transformation of ΔY_(r) to ΔZ in the lossysense circuit (e.g., sense circuit 741). The residual angle error of anexample series resonant configuration of the circuit 540 and for anexample object 110 is provided in TABLE 4.

In an example implementation, the residual error due to thetransformation of ΔY_(r) to ΔZ is reduced by measuring the impedanceZ_(11,0) at two or more substantially different frequencies, supposingabsence of a foreign object, and by determining the elements of anequivalent circuit model (e.g., the equivalent circuit model of FIG. 7K)based on the measured impedances Z_(11,0) employing a best fit method.In some implementations, these two or more frequencies include at leastthe frequency of the minimum and the maximum of |Z_(11,0)(ω)|.

In some implementations and configurations, the change ΔY in theadmittance Y₁₁, if correctly measured at the measurement port 748,directly relates to the reflected admittance ΔY_(r) as previouslydefined. Therefore, in an aspect of object discrimination, the circuit740 may be configured to determine the angle arg{ΔY} and thus the anglearg{ΔY_(r)} with sufficient accuracy. However, in some implementations,measuring the admittance Y₁₁ including the change ΔY may be subject toerrors for various reasons. In particular, there may exist an unknownphase error between the generated sense voltage V₀ as generated by thevoltage source 552 and the current I as measured by the currentmeasurement circuit 550 causing an error in the angle arg{Y₁₁} and thusin the admittance change ΔY related to ΔY_(r).

Analogously to reactance compensation in the sense circuit 701 of FIG.7A, susceptance compensation (resonance tuning) of the sense circuit 741may provide a mean to calibrate the admittance measurement and henceimprove its accuracy e.g., with respect to the angle arg{Y₁₁}. In animplementation configured for parallel resonance, the circuit 740 iscalibrated according to the procedure as previously described withreference to the circuit 540 of FIG. 5C. Nevertheless, a residual errormay remain in the angle arg{ΔY} due to the impact of the seriescapacitor 746. In some implementations, the error in the angle arg{ΔY}or in the angle arg{ΔY,} is reduced analogously to the procedures aspreviously discussed with reference to FIG. 5A.

In an implementation configured for series resonance as defined above,the circuit 740 may be configured to measure the impedance Z₁₁ andcorresponding changes ΔZ of Z₁₁ as caused by the object 110, 112, 114,or vehicle 330. However, as opposed to the parallel resonantconfiguration, the angle arg{ΔZ} may disagree with the angle arg{ΔZ_(r)}of the reflected impedance as previously defined with reference to FIG.7A. There may be a substantial offset between arg{ΔZ} and arg{ΔZ_(r)} asshown below in TABLE 2. Therefore, the phase calibration procedure asdescribed above may not directly apply to the series resonantconfiguration.

In an example implementation configured for series resonance,calibration is performed by measuring the impedance Z_(11,0) atsubstantially different frequencies, supposing absence of a foreignobject, and by determining the elements of an equivalent circuit model(e.g., the equivalent circuit model of FIG. 7K) based on the measuredimpedances Z_(11,0) employing a best fit method. In someimplementations, at least the frequency of the minimum and the maximumof |Z_(11,0) are measured to determine the unknown parameter values ofthe equivalent circuit model.

In an implementation variant of the circuit 740 of FIG. 7E (not shownherein), a ground-based current measurement circuit 550 and anon-ground-based (floating) voltage source 552 is used.

In an implementation variant of the circuit 740 of FIG. 7E (not shownherein), both the voltage source 552 and the current measurement circuit550 are ground-based and a transformer is used for purposes of galvanicseparation. The transformer may be considered inserted between themeasurement port 748 and the series capacitor 746.

The circuit 750 of FIG. 7F illustrates yet another exampleimplementation based on measuring a complex impedance Z₁₁ of a one-portinductive sense circuit 751 (shown in FIG. 7F as the circuit on theright side of the dashed line). More specifically, the impedance Z₁₁ ismeasured at the measurement port 758 (indicated in FIG. 7F by a terminaland a dashed line) by applying, from a voltage source 552, a sinusoidalvoltage V₀ and by measuring, using a current measurement circuit 550,the complex short-circuit current I as previously mentioned withreference to FIG. 5A (voltage source current measurement technique).

The circuit 750 is a modification of the circuit 740 of FIG. 7E tooperate with the double-ended sense electrode 732 of FIG. 7D. Thecircuit 750 includes the sense circuit 751 comprising the double-endedsense electrode 732 having a differential-mode capacitance C, a parallelinductor 754 having an inductance L_(p), a series capacitor 756 having acapacitance C_(s) electrically connected in series to the parallelconnection of the inductor 754 and the double-ended sense electrode 732.In the example implementation as shown by FIG. 7F, the series capacitor756 is split into two capacitors, each with a capacitance 2C_(s),providing a symmetric topology. Further, the sense circuit 751 includesa transformer 757 having a transformation ratio n_(T):1 and a secondaryreferred main inductance L_(m). Its secondary winding is electricallyconnected to the series capacitor 756, while its primary winding iselectrically connected to the measurement port 758. The circuit 750further illustrates the sense signal voltage source 552 and the currentmeasurement circuit 550 both electrically connected to the sense circuit751 at the measurement port 758. As opposed to the sense circuit 741 ofFIG. 7E, the current measurement circuit 550 of the circuit 740 isground-based.

Though not indicated in FIG. 7F for purposes of illustration, thecapacitive and inductive elements of the sense circuit 751 may causeelectrical losses that may be represented by a respective equivalentseries resistance as previously discussed with reference to FIG. 7A and7E.

As previously discussed with reference to the circuit 700 of FIG. 7A,the circuit 750 of FIG. 7F may be configured to be operated at parallelresonance substantially at the nominal sense frequency. Alternatively,it may be configured for series resonance substantially at the nominalsense frequency.

In some implementations, the transformer's 757 main inductance L_(m),the series capacitor 756, and the parallel inductor 754 are used forpurposes of resonance tuning and impedance transformation, e.g., totransform the impedance Z₁₁ to match the sense circuit 711 with anoperating impedance range as previously mentioned with reference toFIG. 1. The inductance ratio L_(m)/L_(p) and the capacitance ratioC/C_(s) may be parameters to control the impedance magnitude |Z_(11,0)|.

In some implementations, the transformer 757 is a 1:1 transformer andserves for balancing. In other implementations, it is a n_(T):1transformer (n_(T)≠1) and is also used for impedance transformation.

In a further aspect, the double-ended sense electrode's 732 capacitanceC in combination with the parallel inductor 754, the series capacitor756, and the transformer's 757 main inductance L_(m) form a higher orderhigh pass filter to attenuate a low frequency disturbance component inthe current I for purposes as previously discussed in connection withFIG. 5A.

With reference to FIG. 1, FIG. 7F also illustrates the objects 110 ,112, and 114 proximate to the double-ended sense electrode 732. Aspreviously discussed with reference to FIG. 1, presence of the object110, 112, 114, or vehicle 330 may cause a change in one or moreelectrical characteristics of the sense circuit 751. As non-limitingexamples, it may a change the capacitance C and an equivalent seriesresistance (not shown in FIG. 7F) resulting in an impedance change ΔZwith respect to the impedance Z_(11,0) as measured in absence of aforeign object with reference to FIG. 3.

In an implementation variant of the circuit 750 of FIG. 7F (not shownherein), the inductor 754 is replaced by the transformer 757 having asecondary referred main inductance L_(m)=L_(p). In this implementationvariant, the series capacitor 756 may be composed of a single capacitor756 having capacitance C_(s) directly connecting to the terminal of themeasurement port 758 and to the transformer's 757 primary winding.

The circuit 760 of FIG. 7G illustrates an example implementation basedon measuring a complex transimpedance Z₂₁ of a two-port capacitive sensecircuit 761 (shown in FIG. 7G as the circuit between the left and theright dashed line). More specifically, the transimpedance Z₂₁ ismeasured by applying, from the current source 512, a sinusoidal currentI_(0,1) at the sense frequency with a defined amplitude and phase to themeasurement port 768 (indicated in FIG. 7G by a terminal and a dashedline) and by measuring, using a voltage measurement circuit 510, thecomplex open-circuit voltage V₂ (amplitude and phase) at the measurementport 769 (indicated in FIG. 7G by terminal 769 and a dashed line) aspreviously described with reference to FIG. 5D.

The sense circuit 761 comprises a double-electrode capacitive senseelement 762 comprising a first single-ended sense electrode 762 a havinga single terminal 763 a and a second single-ended-sense electrode 762 bhaving a single terminal 763 b. The sense circuit 761 further comprisesa first series inductor 764 having an inductance L_(s,1) electricallyconnected in series to the first sense electrode 762 a at the terminal763 a and a second series inductor 765 having an inductance L_(s,2)electrically connected in series to the second sense electrode 762 b atthe terminal 763 b. The sense circuit 761 further comprises a firstparallel inductor 766 having an inductance L_(p,1) electricallyconnected to the first series inductor 764 and in parallel to themeasurement port 768 and a second parallel inductor 767 having aninductance L_(p,2) electrically connected to the second series inductor765 and in parallel to the measurement port 769. The circuit 760 furtherillustrates the sense signal current source 512 connected to themeasurement port 768 and the voltage measurement circuit 510 connectedto the measurement port 769.

Though not indicated in FIG. 7G for purposes of illustration, theinductors 764, 765, 766, and 767 may also cause electrical losses thatmay be represented by a respective equivalent series resistance asindicated in FIG. 7A.

FIG. 7G indicates, in dashed lines, a capacitance C_(ag) between thefirst sense electrode 762 a and ground, a capacitance C_(ab) between thefirst sense electrode 762 a and the second sense electrode 762 b, and acapacitance C_(bg) between the second sense electrode 762 b and ground.The capacitances C_(ag), and C_(bg) may include other capacitances asdiscussed with reference to FIG. 7A.

Analogously to the self-inductances L₁, L₂, and the mutual inductanceL_(M) of a two-port inductive sense element (e.g., inductive senseelement 562 of FIG. 5D comprising the sense coils 562 a and 562 b), afirst self-capacitance C₁, a second self-capacitance C₂, and a mutualcapacitance C_(M) as indicated in FIG. 7G may be attributed to thetwo-port capacitive sense element 762. The self-capacitance C₁ may bedefined as the capacitance as measured between the terminal 763 a of thefirst sense electrode 762 a and ground with the terminal 763 b shortenedto ground. Likewise, the self-capacitance C₂ may be defined as thecapacitance as measured between the terminal 763 b of the second senseelectrode 762 b and ground with the terminal 763 a shortened to ground.FIG. 7G also indicates corresponding equivalent series resistance R₁,R₂, and R_(M) representing electrical losses in the capacitive senseelement 762.

Neglecting any effect of R₁, R₂, and R_(M), the following relations mayapply between the capacitances C_(ag), C_(ab), C_(bg) and thecapacitances C₁, C₂, C_(M):

C ₁ =C _(ag) +C _(ab)  (182)

C ₂ =C _(bg) +C _(ab)  (183)

C_(M)=C_(ab)  (184)

Analogously to the inductive coupling factor, a capacitive couplingfactor may be defined as:

k _(C) =C _(M)(C ₁ C ₂)^(−1/2)  (185)

Substituting C₁, C₂, C_(M) in Equation (185) by Equations (182), (183),and (184) provides:

k _(C) =C _(ab)(C _(ag) +C _(ab))^(−1/2)(C _(bg) +C _(ab))^(−1/2)  (186)

Analogously to the “T”-equivalent circuit model 562-1 of a two-portinductive sense element (e.g., inductive sense element 562) illustratedin FIG. 5I, a two-port capacitive sense element (e.g., capacitive senseelement 762) may be modeled by a “π”-equivalent circuit based oncapacitances C₁, C₂, C_(M) as illustrated in FIG. 7L. Alternatively, atwo-port capacitive sense element (e.g., capacitive sense element 762)may be modeled by an equivalent circuit illustrated by FIG. 7Mcomprising the capacitances C₁, C₂ in parallel to the voltage-controlledcurrent sources with respective currents:

I_(ind,1)=jωC_(M)V₂  (187)

I_(ind,2)=jωC_(M)V₁  (188)

representing the currents induced into the primary and secondaryelectrodes, respectively, as indicated in FIG. 7M. As with theequivalent circuit models 762-1 of FIG. 7L and 562-1 of FIG. 5I, theequivalent circuit model 762-1 of FIG. 7M is electrically dual to theequivalent circuit model 562-2 of FIG. 5J.

In some implementations, the reactance of L_(s,1) substantiallycompensates for the reactance of C₁ providing a local impedance minimum|Z₁₁| (series resonance) substantially at the nominal sense frequency,while the reactance of L_(s,2) substantially compensates for thereactance of C₂ L₂ providing a local impedance minimum |Z₂₂| (seriesresonance) substantially at the nominal sense frequency.

In another implementation, the sense circuit 761 is configured toprovide a local minimum of the admittance magnitude functions |Y₁₁(ω)|and |Y₂₂(ω)| (parallel resonance) substantially at the nominal sensefrequency.

In a further implementation, the sense circuit 761 is configured toprovide a local minimum of the admittance magnitude function |Y₁₁(ω)|(parallel resonance) and a local minimum of the impedance magnitudefunction |Z₂₂(ω)| (series resonance) substantially at the nominal sensefrequency.

In yet another implementation, the sense circuit 761 is configured toprovide a local minimum of the impedance magnitude function |Z₁₁(ω)|(series resonance) and a local minimum of the admittance magnitudefunction |Y₂₂(ω)| (parallel resonance) substantially at the nominalsense frequency.

In implementations configured for primary-side and secondary-side seriesresonance, the reactance of the parallel inductors 766 and 767 issubstantially higher than the impedance magnitudes |Z₁₁| and |Z₂₂|,respectively, of the sense circuit 761 at the nominal sense frequency.

In a further example implementation, at least one of the seriesinductors 764 and 765 is omitted and the sense circuit 761 is operatedas a non-resonant or partially resonant circuit.

In a further aspect, the capacitance C₁ of the first sense electrode 762a in combination with the first parallel inductor 766 form a 2^(nd)order high pass filter to attenuate a low frequency disturbancecomponent in the voltage V₁. Likewise, the capacitance C₂ of the secondsense electrode 762 b form a 2^(nd) order high pass filter to attenuatea low frequency disturbance component in the voltage V2 for purposes aspreviously discussed in connection with FIG. 5A.

With reference to FIG. 1, the sense circuit 761, the capacitive senseelement 762 (including sense electrodes 762 a and 762 b), and therespective series inductors 764 and 765 may correspond e.g., to thecapacitive sense circuit 108 a, the capacitive sense element 109 a(including a double-ended sense electrode), and the respectiveassociated capacitive elements.

Though not shown herein, other transimpedance measurement techniquessuch as the voltage source current measurement technique or any othercombination may apply (e.g., a voltage source voltage measurementtechnique). In some implementations (also not shown herein), at leastone of the impedances Z₁₁ and Z₂₂ of the sense circuit 781 isadditionally measured to the transimpedance Z₂₁ (e.g., using one or moreof the techniques as previously discussed with reference to FIG. 5A). Inthese alternative implementations, presence of an object (e.g., object114) is determined based on a change in at least one of an impedanceZ₁₁, Z₂₂, and Z_(21.)

Moreover, at least one of an impedance transformation and balancing mayapply to at least one of the primary-side and secondary-side of thesense circuit 761 (not shown herein). More specifically, with referenceto the sense circuit 721 of FIG. 7C, a transformer (e.g., transformer726) may be used instead of the parallel inductors 766 and 767.Alternatively, with reference to the sense circuit 741 of FIG. 7E, aseries capacitor and an inductor connected in parallel to the senseelectrode (e.g., series capacitors 746 and parallel inductor 744) mayapply at least on the primary side.

With reference to FIG. 1, FIG. 7G also illustrates the objects 110, 112,and 114 proximate to the capacitive sense element 762. As previouslydiscussed with reference to FIG. 1, presence of the object 110, 112,114, or vehicle 330 may cause a change in one or more electricalcharacteristics of the sense circuit 761. As non-limiting examples, itmay change at least one of the self-capacitances C₁, C₂, the equivalentseries resistances R₁, R₂, the mutual capacitance C_(M) and the mutualequivalent series resistance R_(M) generally resulting in a change ΔZ ofthe transimpedance Z_(21,0) as measured in absence of a foreign object.Presence of an object (e.g., object 114) may be determined if ΔZsatisfies certain criteria (e.g., the magnitude of ΔZ exceeds adetection threshold) as previously discussed with reference to FIG. 7A.

Using a quasi-ideal current source 512, a change ΔZ in thetransimpedance Z₂₁ (e.g., due to presence of the object 114) manifestsin a change ΔV in the voltage V₂ while the current I_(0,1) remainssubstantially unaffected. Therefore, measuring the complex voltage V₂may be equivalent to measuring the complex transimpedance Z₂₁. In otherwords, the complex voltage V₂ may be indicative of the complextransimpedance Z₂₁ and there may be no requirement for additionallymeasuring the current I_(0,1) thus reducing complexity of themeasurement circuit (e.g., measurement circuit 104 of FIG. 1)

As with the sense circuit 561 of FIG. 5D, the fractional change ΔZ′ (orΔY′) caused by a defined test object (e.g., object 112) placed at adefined position relative to the capacitive sense element 762 may relateto the detection sensitivity of an object detection circuit (e.g., themulti-purpose detection circuit 100 of FIG. 1) based on a two-portcapacitive sense circuit 761. Increasing the fractional change ΔZ′ (orΔY′) may increase a detection sensitivity of the circuit 760. Morespecifically, it may increase a signal-to-noise ratio (SNR) e.g., asdefined by Equation (21)

As non-limiting examples, the fractional change ΔZ′ (or ΔY′) may beincreased by optimizing the design and the arrangement of the senseelectrodes 762 a and 762 b, their integration into the wireless powertransfer structure (e.g., wireless power transfer structure 200 withreference to FIGS. 2 and 3), by resonance tuning using series inductors764 and 765 as previously described, and by improving a Q-factor of thesense circuit 761.

The circuit 770 of FIG. 7H illustrates another example implementationbased on measuring a complex transimpedance Z₂₁ of a two-port capacitivesense circuit 771 (shown in FIG. 7H as the circuit between the left andthe right dashed line). More specifically, the transimpedance Z₂₁ ismeasured by applying, from the current source 512, a sinusoidal currentI_(0,1) at the sense frequency with a defined amplitude and phase to themeasurement port 778 (indicated in FIG. 7H by a terminal and a dashedline) and by measuring, using the voltage measurement circuit 510, thecomplex open-circuit voltage V₂ (amplitude and phase) at the measurementport 779 (indicated in FIG. 7H by a terminal and a dashed line) aspreviously described with reference to FIG. 5D.

As opposed to the sense circuit 761, the sense circuit 771 is operatedin a differential mode. It may be split into a first branch and a secondbranch with an equal topology. The sense circuit 771 may besubstantially symmetric (balanced) with respect to its capacitances andinductances. The sense circuit 771 includes a quad-electrode capacitivesense element 772 comprising a first double-ended sense electrodecomposed of sense electrodes 772 a and 772 b having respective terminals773 a and 773 b and a second double-ended sense electrode composed ofsense electrodes 772 c and 772 d having respective terminals 773 c and773 d. The first double-ended sense electrode 772 a/b is electricallyconnected to a differential-mode inductor 774 providing an inductanceL_(a,1)/2 in each branch. The second double-ended sense electrode 772c/d is electrically connected to a differential-mode inductor 775providing an inductance L_(s,2)/2 in each branch. The sense circuit 771also includes a transformer 776 with a transformation ratio n₁:1 and asecondary referred main inductance L_(m,1). Its primary winding iselectrically connected in parallel to the measurement port 778, whileits secondary winding is electrically connected to the differential-modeinductor 774. Further, the sense circuit 771 includes a transformer 777with a transformation ratio 1:n₂ and a primary referred main inductanceL_(m,2). Its primary winding is electrically connected to thedifferential-mode inductor 775, while its secondary winding iselectrically connected in parallel to the measurement port 779. Thecircuit 770 further illustrates the sense signal current source 512electrically connected to the measurement port 778 and the voltagemeasurement circuit 510 electrically connected to the measurement port779.

Though not indicated in FIG. 7H for purposes of illustration, the sensecircuit 771 may cause electrical losses in the inductive and capacitiveelements that may be represented by a respective equivalent seriesresistance as previously described with reference to FIGS. 7A and 7G.More specifically, the capacitive sense element 772 may cause electricallosses that may be represented by corresponding equivalent seriesresistances R₁, R₂, R_(M).

Further, FIG. 7H indicates, in dashed lines, a plurality of capacitancescomprising a capacitance C_(ab) between electrodes 772 a and 772 b, acapacitance C_(ac) between electrodes 772 a and 772 c, a capacitanceC_(ad) between electrodes 772 a and 772 d, a capacitance C_(bc) betweenelectrodes 772 b and 772 c, a capacitance C_(bd) between electrodes 772b and 772 d, and a capacitance C_(cd) between electrodes 772 c and 772d. These capacitances may refer to capacitances as measured betweenrespective terminals 773 a, 773 b, 773 c, and 773 d with thedifferential-mode inductors 774 and 775 disconnected from the respectivesense electrodes. The capacitances C_(ab) and C_(cd) may further includevarious capacitances as previously illustrated with reference to FIG.7A.

As with the capacitive sense element 762 of FIG. 7G, a firstself-capacitance C₁, a second self-capacitance C₂, and a mutualcapacitance C_(M) as indicated in FIG. 7H may be attributed to thecapacitive sense element 772. Since the sense circuit 771 is operated ina differential mode, these capacitances may be considered asdifferential-mode capacitances. Analogously to the circuit 760 of FIG.7G, the quad-electrode capacitive sense element (e.g., capacitive senseelement 762) may be modeled by the “π”-equivalent circuit model based oncapacitances C₁, C₂, C_(M) as illustrated in FIG. 7L. If required, theground-based “π” circuit may be replaced by an equivalentground-symmetric network having a mutual capacitance C_(M)/2 in eachbranch (not shown herein). Alternatively, it may be modeled by theequivalent circuit model illustrated in FIG. 7M comprising thecapacitances C₁, C₂ in parallel to the voltage-controlled currentsources.

The following relations between the capacitances C_(ab), C_(ac), C_(ad),C_(bc), C_(bd), C_(cd) and C₁, C₂, C_(M) may be found:

C ₁ =C _(ab)+((C _(ac) +C _(ad)) (C _(bc) +C _(bd))/93 C)  (189)

C ₂ =C _(cd)+((C _(ac) +C _(bc))(C _(ad) +C _(bd))/93 C)  (190)

C _(M)=(C _(ac) C _(bd) −C _(ad) C _(bc))/ΣC  (191)

with ΣC denoting the sum of the coupling capacitances:

ΣC=(C _(ac) +C _(ad) +C _(bc) +C _(bd))  (192)

For an entirely symmetric capacitive sense element 772 withcapacitances:

C_(ab)=C_(cd)=C_(a)  (193)

C_(ac)=C_(db)=C_(b)  (194)

C_(ad)=C_(bc)=C_(c)  (195)

the mutual capacitance of Equation (191) becomes:

C _(M)=(C _(b) −C _(c))/2  (196)

and the self-capacitances of Equations (189) and (190):

C ₁ =C ₂ =C _(a)+((C _(b) +C _(c))/2)  (197)

and the capacitive coupling factor:

k _(C)=(C _(b) −C _(c))/(2C _(a) +C _(b) +C _(c))  (198)

In some implementations, the reactance of L_(s,1) substantiallycompensates for the reactance of C₁ providing a local impedance minimum|Z₁₁| (series resonance) substantially at the nominal sense frequency,while the reactance of L_(s,2) substantially compensates for thereactance of C₂ providing a local impedance minimum |Z₂₂| (seriesresonance) substantially at the nominal sense frequency.

In another implementation, the sense circuit 771 is configured toprovide a local minimum of the admittance magnitude functions |Y₁₁(ω)|and |Y₂₂(ω)| (parallel resonance) substantially at the nominal sensefrequency.

In a further implementation, the sense circuit 771 is configured toprovide a local minimum of the admittance magnitude function |Y₁₁(ω)|(parallel resonance) and a local minimum of the impedance magnitudefunction |Z₂₂(ω)| (series resonance) substantially at the nominal sensefrequency.

In yet another implementation, the sense circuit 771 is configured toprovide a local minimum of the impedance magnitude function |Z₁₁(ω)|(series resonance) and a local minimum of the admittance magnitudefunction |Y₂₂(ω)| (parallel resonance) substantially at the nominalsense frequency.

In implementations configured for primary-side and secondary-side seriesresonance, the reactance of the transformer's 776 primary referred maininductance and the transformer's 776 secondary referred main inductanceis substantially higher than the impedance magnitudes |Z₁₁| and |Z₂₂|,respectively, of the sense circuit 771 at the nominal sense frequency.

In a further example implementation, at least one of thedifferential-mode inductors 774 and 775 is omitted and the circuit 771is operated as a non-resonant or partially resonant circuit.

In a further aspect, the capacitance C₁ of the first double-ended senseelectrode 772 a/b in combination with the first transformer's 776 maininductance L_(m,1) form a 2^(nd) order high pass filter to attenuate alow frequency disturbance component in the voltage V₁. Likewise, thecapacitance C₂ of the second double-ended sense electrode 772 c/d incombination with the second transformer's 777 main inductance L_(m,2)form a 2^(nd) order high pass filter to attenuate a low frequencydisturbance component in the voltage V₂ for purposes as previouslydiscussed in connection with FIG. 5A.

With reference to FIG. 1, the sense circuit 771, the capacitive senseelement 772 (including electrodes 772 a, 772 b, 772 c, and 772 d), andthe respective differential-mode inductors 774 and 775 may corresponde.g., to the capacitive sense circuit 108 a, the capacitive senseelement 109 a (including a pair of double-ended sense electrodes, notshown in FIG. 1), and the respective associated capacitive elements.

Though not shown herein, other transimpedance and impedance measurementtechniques as previously discussed with reference to FIG. 7G may apply.

With reference to FIG. 1, FIG. 7H also illustrates the objects 110, 112,and 114 proximate to the capacitive sense element 772. As previouslydiscussed with reference to FIG. 7G, presence of the object 110, 112,114, or vehicle 330 may cause a change in one or more electricalcharacteristics of the sense circuit 771. As non-limiting examples, itmay change at least one of the capacitances C₁, C₂, C_(M) and theequivalent series resistances R₁, R₂, R_(M) generally resulting in achange ΔZ of the transimpedance Z_(21,0) as measured in absence of aforeign object. Presence of an object (e.g., object 114) may bedetermined as previously discussed with reference to FIG. 7A.

As with the sense circuit 561 of FIG. 5D, the fractional change ΔZ′ (orΔY′) caused by a defined test object (e.g., object 112) placed at adefined position relative to the capacitive sense element 772 may relateto the detection sensitivity of an object detection circuit (e.g., themulti-purpose detection circuit 100 of FIG. 1) based on a two-portcapacitive sense circuit 771. Increasing the fractional change ΔZ′ (orΔY′) may increase a detection sensitivity of the circuit 770 aspreviously discussed with reference to FIG. 7G.

As non-limiting examples, the fractional change ΔZ′ (or ΔY′) may beincreased by optimizing the design and the arrangement of thedouble-ended sense electrodes 772 a/b and 772 c/d, their integrationinto the wireless power transfer structure (e.g., wireless powertransfer structure 200 with reference to FIGS. 2 and 3), by resonancetuning using series inductors (e.g., differential-mode inductors 774 and775) as previously described, and by improving a Q-factor of the sensecircuit 761.

In an example implementation and in analogy to the implementation of theinductive sense element 562 as described with reference to FIG. 5D, thecapacitive sense element 772 is configured for the mutual capacitanceC_(M) to vanish in absence of a foreign object, resulting in atransimpedance |Z_(21,0)| that is substantially zero. According toEquation (191), the mutual capacitance vanishes (C_(M)≈0) if thecapacitances C_(ab), C_(cd), C_(ad), and C_(bc) satisfy:

C_(ac)C_(bd)≈C_(ad)C_(bc)  (199)

For an entirely symmetric capacitive sense element 772 using Equation(196), the mutual capacitance vanishes if:

C_(b)≈C_(c)  (200)

An example implementation of a capacitive sense element 772 using anarrangement of four single-ended sense electrodes that may provide asubstantially zero mutual capacitance (C_(M)≈0) is illustrated in FIG.12A.

The circuit 780 of FIG. 7I illustrates yet another exampleimplementation based on measuring a complex transimpedance Z₂₁ of atwo-port capacitive sense circuit 781 (shown in FIG. 7I as the circuitbetween the left and the right dashed line). More specifically, thetransimpedance Z₂₁ is measured by applying, from the voltage source 552,a sinusoidal voltage V_(0,1) at the sense frequency with a definedamplitude and phase to the measurement port 788 (indicated in FIG. 7I bya terminal and a dashed line) and by measuring, using the currentmeasurement circuit 550, the complex short-circuit current I₂ (amplitudeand phase) at the measurement port 789 (indicated in FIG. 7I by terminal789 and a dashed line). The circuit 780 of FIG. 7I may be considered aselectrically dual to the circuit 580 of FIG. 5E.

The sense circuit 781 includes the double-electrode capacitive senseelement 762 with reference to FIG. 7G composed of the first single-endedsense electrode 762 a and the second single-ended sense electrode 762 bwith reference to FIG. 7G. FIG. 7I also indicates capacitances C_(ag),C_(ab), C_(bg) and the related capacitances C₁, C₂, C_(M) as previouslydiscussed with reference to FIG. 7G. The first electrode 762 a iselectrically connected to the first terminal of a parallel inductor 784having an inductance L_(p) and the second electrode 762 b iselectrically connected to the second terminal of the parallel inductor784. Further, the first electrode 762 a and the second electrode 762 bis electrically connected to the first terminal of a first seriescapacitor 786 and to the first terminal of a second series capacitor787, respectively, while the second terminal of the series capacitor 786and 787 is electrically connected to the measurement ports 788 and 789,respectively.

Though not indicated in FIG. 7I for purposes of illustration, the sensecircuit 781 may cause electrical losses in the inductive and capacitiveelements that may be represented by a respective equivalent seriesresistance as previously described with reference to FIGS. 7A and 7G.More specifically, the capacitive sense element 762 may include theequivalent series resistances R₁, R₂, and R_(M).

The basic topology of the circuit 780 of FIG. 7I equals the topology ofthe circuit 760 of FIG. 7F if the transformer 757 is omitted. Therefore,some implementations or configurations of the circuit 780 measuring atransimpedance Z₂₁ may be considered equivalent to the circuit 760measuring an impedance Z₁₁.

In an example implementation, the sense electrodes 762 a and 762 b aretightly coupled resulting in a capacitive coupling factor kc as definedby Equation (186) that is near unity (k_(C)≈<1).

The sense circuit 781 may be configured to provide a local minimum inthe transimpedance magnitude function |Z_(21,0)(ω)| (series resonance)substantially at a nominal sense frequency. Alternatively, the sensecircuit 581 may be configured to provide a local minimum in thetransadmittance magnitude function |Y_(21,0)(ω)| substantially at thenominal sense frequency.

In an example parallel resonant configuration of the sense circuit 781using a capacitive sense element 762 with k_(C)≈<1, the susceptance ofthe parallel inductor 784 substantially compensates for the susceptanceof the mutual capacitance C_(M) providing a local minimum in thetransadmittance magnitude function |Y_(11,0)(ω)| (parallel resonance)substantially at the nominal sense frequency. The principle of mutualsusceptance compensation may become more evident by contemplating FIG.7L illustrating the “π”-equivalent circuit model 762-1 of the two-portcapacitive sense element 762 and by considering the inductance L_(p) ofthe parallel inductor 784 inserted in parallel to the mutual capacitanceC_(M). With k_(C)≈<1, both the parallel capacitances C₁-C_(M) andC₂-C_(M) become substantially zero.

In this parallel resonant configuration, the capacitances C_(s,1) andC_(s,2) of the series capacitor 786 and 787, respectively, may besimilar or larger than the capacitance C₁ and C₂ of the sense electrodes762 a and 762 b, respectively. Stated in other terms, the admittancemagnitude of the series capacitor 786 and 787 may be substantiallyhigher than the admittance magnitude |Y₁₁| and |Y₂₂|, respectively, ofthe sense circuit 781 at the nominal sense frequency. In thisconfiguration, the series capacitor 786 and 787 may exert a negligibleimpact on the admittances and transadmittance |Y₁₁|, |Y₂₂|, and |Y₂₁|,respectively, at the nominal sense frequency.

In an example series resonant configuration of the sense circuit 781using a capacitive sense element 762 with k_(L)≈<1, the susceptance ofthe parallel inductor 784 overcompensates for the susceptance of themutual capacitance C_(M) at the nominal sense frequency. The residualinductive reactance of the parallel connection of the parallel inductor784 and the mutual capacitance C_(M) is substantially compensated for bythe reactance of the series capacitors 786 and 787 providing atransimpedance Z_(21,0) that is substantially real (resistive).

In this series resonant configuration, the capacitances C_(s,1) andC_(s,2) of the series capacitors 786 and 787, respectively, may besmaller, similar, or larger than the capacitances C₁ and C₂ of the senseelectrodes 762 a and 762 b, respectively. Stated in other terms, theimpedance magnitude of each of the series capacitors 786 and 787 may besubstantially (e.g., 20 times) higher than the impedance magnitudes|Z₁₁| and |Z₂₂|, respectively, as presented at the nominal sensefrequency. In this configuration, the series capacitors 786 and 787exert a significant impact on the impedance and transimpedancemagnitudes |Z₁₁|, |Z₂₂|, and |Z₂₁|, respectively, at the nominal sensefrequency.

In a further aspect, the capacitance C₁ of the first sense electrode 762a in combination with the first series capacitor 786 form a high passfilter to attenuate a low frequency disturbance component in the currentI₁. Likewise, the capacitance C₂ of the second sense electrode 762 b incombination with the second series capacitor 787 form a high pass filterto attenuate a low frequency disturbance component in the current I₂ forpurposes as previously discussed in connection with FIG. 5A.

In yet a further aspect, the capacitance C_(M) of the capacitive senseelement 762 in combination with the parallel inductor 784 form a 2^(nd)order high pass filter to attenuate a differential low frequencydisturbance voltage between the sense electrodes 762 a and 762 b.

With reference to FIG. 1, the sense circuit 781, the capacitive senseelement 762 (including sense electrodes 762 a and 762 b), and theparallel inductor 784 may correspond e.g., to the capacitive sensecircuit 108 a, the capacitive sense element 109 a (including adouble-ended sense electrode), and the respective associated inductiveelement, respectively.

Though not shown herein, other transimpedance and impedance measurementtechniques may apply as previously mentioned with reference to FIG. 7G.

Moreover, as previously mentioned with reference to FIG. 7G, at leastone of an impedance transformation and balancing may apply to at leastone of a primary-side and secondary side of the sense circuit 781 (notshown herein).

With reference to FIG. 1, FIG. 7I also illustrates the objects 110, 112,and 114 proximate to the capacitive sense element 762. As previouslydiscussed with reference to FIG. 1, presence of the object 110, 112,114, or vehicle 330 may cause a change in one or more electricalcharacteristics of the sense circuit 781 as previously discussed withreference to FIG. 7D. Particularly, a change in the mutual capacitanceC_(M), and the equivalent mutual resistance R_(M) generally results in achange ΔZ with respect to the transimpedance Z_(21,0) as measured inabsence of a foreign object.

As with the sense circuit 561 of FIG. 5D, the fractional change ΔZ′ (orΔY′) caused by a defined test object (e.g., object 112) may relate tothe detection sensitivity of the sense circuit 781. As non-limitingexamples, the fractional change ΔZ′ (or ΔY′) may be increased byoptimizing the design and the arrangement of the sense electrodes 762 aand 762 b, their integration into the wireless power transfer structure(e.g., wireless power transfer structure 200 with reference to FIGS. 2and 3), by resonance tuning using the parallel inductor 784 aspreviously described, and by improving a Q-factor of the sense circuit781.

It may be appreciated that the sense circuit 781 configured for acapacitive coupling factor k_(C)≈<1 reduces the impact of the equivalentseries resistances R₁ and R₂ on the fractional change, if compared e.g.,to the circuit 700 of FIG. 7A. Moreover, the impedance change ΔZ mayreflect electrical properties of an object (e.g., object 110) asdiscussed with reference to the circuit 700 of FIG. 7A. It may furtherallow calibration and correction of the angle arg{ΔZ} in a procedure aspreviously described with reference to FIG. 5A.

FIGS. 7J and 7K illustrate equivalent circuit models 700-1 and 740-1,respectively, used below for purposes of a theoretical analysis andperformance comparison. More specifically, the equivalent circuit model700-1 is used to analyze the circuit 700 of FIG. 7A, the circuit 710 ofFIG. 7B (using the parallel capacitor 715), and the circuit 720 of FIG.7C (using the transformer 726) , while the equivalent circuit model740-1 serves for the analysis of the circuit 740 of FIG. 7E (theelectrically dual of the circuit 700). Each of the circuits 700, 710,720, and 740 are analyzed with respect to its series and parallelresonant configuration and with respect to various characteristics suchas the impedance and the Q-factor of the sense circuit at resonance, thefractional change, and the various SNRs as previously defined withreference to FIGS. 5F and 5G.

For purposes of comparison, an identical sense electrode 702 an equalsense electrode current level |I_(C)| is assumed for both configurationsof the circuits 700, 710, 720, and 740, though practical implementationsconfigured for parallel resonance may prefer a sense electrode 702 witha higher capacitance C. Comparing SNRs at the same sense electrodecurrent level |I_(C)| may be meaningful e.g., if the current level|I_(C)| is emission or power constraint. Further, it is assumed that thecircuits in both configurations are adjusted to a common resonantfrequency substantially corresponding with the nominal sense frequencythat is substantially higher than the WPT operating frequency.

The equivalent circuit model 700-1 as illustrated in FIG. 7J comprisesthe sense electrode's 702 capacitance C, the series inductor's 704inductance L_(s) and its equivalent series resistance R_(Ls), theparallel inductor's 706 inductance L_(p) and its equivalent seriesresistance R_(Lp), an ideal sense signal current source 512 and an idealvoltage measurement circuit 510. As already noted above, FIG. 7J alsoincludes the capacitance C_(p) of the parallel capacitor 715 (asoptional in dashed lines) with reference to the circuit 710 of FIG. 7B.It may be appreciated that in practical implementations losses incapacitors (including the sense electrode 702) are generallysubstantially lower than losses in inductors. Therefore, an equivalentseries resistances R and R_(Cp) of the sense electrode 702 and theoptional parallel capacitor 715, respectively, (both not shown in FIG.7J) are neglected in the analysis below. The equivalent circuit model700-1 further includes an impedance ΔZ_(r) in series to the capacitanceC representing the reflected impedance of the object 110, 112, or 114proximate to the sense electrode 702 as previously discussed withreference to FIG. 7A. (The reflected impedance ΔZ_(r) may be regarded asthe object 110, 112, or 114 as illustrated in FIG. 7A abstracted away).The equivalent circuit model 700-1 also includes a noise voltage sourceV_(sn) in series to the capacitance C representing the noise voltageinduced into the sense electrode 702 primarily by the electric field asgenerated when WPT is active. The noise voltage V_(sn) may include anylow frequency component (e.g., at the fundamental of the WPT operatingfrequency and harmonics thereof) as well as any high frequency component(e.g., switching noise at the sense frequency). The equivalent circuitmodel 700-1 of the circuit 700 further indicates the impedance Z₁₁ andthe admittance Y₁₁ (=1/Z₁₁), the sense signal current I₀ with anadditive noise component I_(0,n), the sense signal voltage V with anadditive noise voltage V_(n), and the measurement port 708 (indicated bythe terminal and the dashed line) where the current I₀+I_(0,m) isinjected, the voltage V+V_(n) is measured, and where Z₁₁ or Y₁₁ referto.

It may be appreciated that the equations for the series and parallelresonant configuration of the sense circuit 701 with respect to theimpedance Z₁₁, the admittance Y₁₁, the respective resonant angularfrequencies ω_(s) and ω_(p), the impedance change ΔZ, the admittancechange ΔY, the fractional changes ΔZ′ and ΔY′, and the various circuitintrinsic and extrinsic SNRs as previously defined with reference toFIG. 5F may be derived analogously to Equations (27) to (177) byreplacing C_(s) by C, L by L_(s), R by R_(Ls), etc. Therefore, themathematical derivations are generally omitted. Because the equivalentcircuit model 700-1 applies to the circuit 700 of FIG. 7A or the circuit720 of FIG. 7C, the reference numerals 700 and 720, respectively, areused instead in the following theoretical analysis.

To analyze the series and parallel resonant configuration of the circuit700 of FIG. 7J, the common assumptions:

ωL_(s)>>R_(Ls)  (201)

ωL_(p)>>R_(Lp)  (202)

|ΔZ _(r) |<<R  (203)

are made for a frequency range about the resonant frequency.

In an implementation configured for series resonance and with areactance:

ωL _(p) >>|Z ₁₁|  (204)

in a frequency range about the resonant frequency, the impedance Z₁₁ atthe measurement port 708 of the circuit 700 of FIG. 7J in presence of anobject (e.g., object 112) may be expressed as:

Z ₁₁ ≈R _(Ls) +jωL _(s)+(jωC)⁻¹ +ΔZ _(r)  (205)

In absence of a foreign object, a local minimum of |Z_(11,0)(ω)| (seriesresonance) occurs substantially at an angular frequency ω satisfying:

(jωL _(s))⁻¹ +jωC≈0  (206)

yielding the series resonant angular frequency:

ω_(s)≈(CL _(s))^(−1/2)  (207)

At this frequency, the impedance Z_(11,0) becomes substantially real:

Z _(11,0) ≈Re{Z _(11,0) }=R _(s) ≈R _(Ls)  (208)

with R_(s) denoting the series resonant resistance, while the impedanceZ₁₁ in presence of an object (e.g., object 112) is approximately:

Z ₁₁ ≈R _(s) +ΔZ≈R _(Ls) +ΔZ _(r)  (209)

with ΔZ_(r) referring to the reflected impedance as previously definedwith reference to FIG. 7A.

The fractional change ΔZ′ for the series resonant configuration of thecircuit 700 of FIG. 7J becomes approximately:

ΔZ′=ΔZ/R _(s) ≈ΔZ _(r) /R _(Ls)  (110)

Defining the normalized reflected impedance:

ΔZ_(r)′=ΔZ_(r)ω_(s)C  (211)

the Q-factor of the series inductor 704:

Q _(Ls)=ω_(s) L _(s) /R _(Ls)  (212)

and the Q-factor of the series resonant configuration of the sensecircuit 701 of FIG. 7J:

Q _(s)≈1/(ω_(s) CR _(s))≈Q _(Ls)  (213)

the fractional change may also be written in terms of ΔZ_(r)′ and Q_(s)or Q_(Ls):

ΔZ′≈Q_(s)ΔZ_(r)′≈Q_(Ls)ΔZ_(r)′  (214)

To analyze the parallel resonant configuration of the equivalent circuitmodel 700-1 of FIG. 7J, the following additional assumption:

|ωL _(s)−(ωC)⁻¹ |>>R _(Ls)  (215)

is made for a frequency range about the resonant frequency. Theadmittance Y₁₁ at the measurement port 708 in presence of an object(e.g., object 112) may be expressed:

Y ₁₁=(R _(Lp) +jωL _(p))⁻¹+(R _(Ls) +jωL _(s)+(jωC)⁻¹ +ΔZ _(r))⁻¹  (216)

and using Equation (38) approximated as:

Y ₁₁≈(jωL _(p))⁻¹ −R _(Lp)(jωL _(p))⁻²+(jωL _(s)+(jωC)⁻¹)⁻¹−(R _(Ls) +ΔZ_(r)) (ωL _(s)+(ωC)⁻¹)⁻²  (217)

In absence of a foreign object, a local minimum of |Y_(11,0)(ω)|(parallel resonance) occurs substantially at an angular frequency ωsatisfying:

(jωC)⁻¹+jω(L_(s)+L_(p))  (218)

yielding for the parallel resonant angular frequency:

ω_(p)≈(C(L _(s) +L _(p)))^(−1/2)  (219)

At this frequency, the admittance Y_(11,0) becomes substantially real:

Y _(11,0) ≈Re{Y _(11,0) }=G _(p)≈(R _(Ls) +R _(Lp))/(ω_(p) L_(p))²  (220)

with G_(p) denoting the parallel resonant conductance, while theadmittance Y₁₁ in presence of an object (e.g., object 112) isapproximately:

Y ₁₁ ≈G _(p) +ΔY≈(R _(Ls) +R _(Lp) +ΔZ _(r))/(ω_(p) L _(p))²  (221)

with:

Δ≈ΔZ _(r)/(ω_(p) L _(p))²  (222)

Defining the Q-factor of the series inductor 704 in terms of L_(s) andR_(Ls):

Q _(Ls)≈ω_(p) L _(s) /R _(Ls)  (223)

the Q-factor of the parallel inductor 706 in terms of L_(p) and R_(Lp):

Q _(Lp)≈ω_(p) L _(p) /R _(Lp)  (224)

and the inductance ratio:

n _(L) =L _(s) /L _(p)  (225)

the admittance Y_(11,0) at ω_(p) may be expressed as:

Y _(11,0) =G _(p)≈(1+n _(L))((1/Q _(Lp))+(n _(L) /Q _(Ls))ω_(p) C  (226)

For Q_(Lp)=Q_(Ls) and n_(L)>1, the parallel resonant conductance G_(p)as presented at the measurement port 708 becomes approximately(1+n_(L))² ω_(p) C/Q_(Ls). Using Equations (219) and (225), theadmittance Y₁₁ at ω_(p) of Equations (221) may also be expressed as:

Y ₁₁ ≈G _(p) +ΔY≈G _(p)+(1+n _(L))²(ω_(p) C)²  (227)

According to Equation (226), the admittance Y₁₁ at ω_(p) of Equation(225) of the sense circuit 701 of FIG. 7J can be increased by increasingthe inductance ratio n_(L)=L_(s)/L_(p), while maintaining parallelresonance substantially at the nominal sense frequency. Therefore, insome implementations, the parallel resonant configuration of the circuit700 of FIG. 7A is employed as an alternative to using a transformer(e.g., transformer 726) for transforming the admittance Y₁₁ to be withina suitable operating admittance range as previously discussed withreference to FIGS. 7B and 7C.

Based on Equation (9), the fractional change ΔY′ for the parallelresonant configuration of the circuit 700 of FIG. 7J may be written as:

ΔY′=ΔY/G _(p) ≈ΔZ _(r)/(R _(Ls) +R _(Lp))  (228)

showing that the admittance change ΔY is substantially proportional tothe reflected impedance ΔZ_(r). Therefore, the angle arg{ΔY} of themeasured admittance change ΔY is indicative of the angle arg{ΔZ_(r)}. Insome implementations, the accuracy of the measured angle is improved byapplying the angle correction based on the calibration procedure aspreviously described with reference to the circuit 500 of FIG. 5A.Defining the Q-factor of the parallel resonant configuration of thesense circuit 701 of FIG. 7J as:

Q _(p)≈ω_(p)(L _(p) +L _(s))/(R _(Ls) +R _(Lp))  (229)

which may be expresses in terms of the Q-factors:

Q _(p) ≈Q _(Ls)(1+n _(L))/((Q _(Ls) /Q _(Lp))+n _(L))  (230)

and the normalized reflected impedance:

ΔZ_(r)′=ΔZ_(r)ω_(p)C  (231)

the fractional change ΔY′ may also be written as:

ΔY′≈Q_(p)ΔZ_(r)′  (232)

With reference to the circuit 710 of FIG. 7B, adding the parallelcapacitor 715 to the circuit 700 of FIG. 7J, the impedance Z₁₁ atmeasurement port 708 in presence of an object (e.g., object 112) for theseries resonant configuration becomes:

Z ₁₁ ≈R _(Ls) +jωL _(s)+(jωC _(p)+((jωC)⁻¹ +ΔZ _(r))⁻¹)⁻¹  (233)

Using the approximation of Equation (38), the impedance Z₁₁ may beapproximated as:

Z ₁₁ ≈R _(Ls) +jωL _(s)+(jω(C+C _(p)))⁻¹ +ΔZ _(r) C ²/(C+C _(p))²  (234)

Series resonance in absence of a foreign object occurs approximately atan angular frequency satisfying:

(jω(C+C _(p)))⁻¹ +jωL _(s)≈0  (235)

yielding the series resonant angular frequency:

ω_(s)≈(L _(s)(C+C _(p))^(−1/2)  (236)

At this frequency, the impedance Z_(11,0) becomes substantially real:

Z _(11,0) ≈Re{Z _(11,0) }=R _(s) ≈R _(Ls)  (237)

with R_(s) denoting the series resonant resistance, while the impedanceZ₁₁ in presence of an object (e.g., object 110) is approximately:

Z ₁₁ ≈R _(s) +ΔZ≈R _(Ls) +ΔZ _(r) C ²/(C+C _(p))²  (238)

Defining the capacitance ratio:

n _(C) =C/C _(p)  (239)

the impedance Z_(11,0) of Equation (238) may also be expressed in termsof the Q-factor C_(Ls) of the series inductor 714, the capacitance C ofthe sense electrode 702, and the capacitance ratio n_(C) as:

Z _(11,0) ≈R _(s) ≈n _(C)/((1+n _(C))ω_(s) CQ _(Ls))  (240)

According to Equation (240), the impedance Z₁₁ of the sense circuit 711of FIG. 7B can be reduced by decreasing the capacitance ration_(C)=C/C_(p), while maintaining the series resonance ω_(s)substantially at the nominal sense frequency. For n_(C)=½, the seriesresonant resistance R_(s) amounts to ⅓ of that of the sense circuit 501of FIG. 5F. Therefore, in some implementations, the series resonantconfiguration of the circuit 710 of FIG. 7B using the parallel capacitoris employed as an alternative to using a transformer (e.g., transformer726 of FIG. 7C) for transforming the impedance Z₁₁ to be within asuitable operating admittance range as discussed with reference to FIG.7C.

Based on Equation (238), the impedance change ΔZ resulting at themeasurement port 718 of the sense circuit 711 of FIG. 7B becomesapproximately:

ΔZ≈(n _(C)/(1+n _(C)))² ΔZ _(r)  (241)

Defining the Q-factor of the series resonant configuration of the sensecircuit 711 of FIG. 7B as:

Q _(s)≈ω_(s) L _(s) /R _(s) =Q _(Ls)  (242)

the fractional change ΔZ′ may be expressed as:

ΔZ′≈ΔZ/R _(s) ≈n _(C)/(1+n _(C))Q _(s) ΔZ _(r)′  (243)

Based on Equation (243), adding a parallel capacitor 715 (as shown inFIG. 5C) for purposes of impedance matching as previously discussed withreference to FIGS. 5A and 5B may substantially reduce the fractionalchange. Though not shown herein, the same reduction may also apply tothe parallel resonant configuration of the circuit 710 of FIG. 7B.

In some implementations based on the circuit 700 of FIG. 7J, the currentlevel I₀ of the current source 512 is adjusted to match a specifiedcurrent |I_(C)| in the sense electrode 702. Therefore, in an aspect, therequired current level I₀, the resulting voltage V at the measurementport 708, and the drive power level P may be considered. For the seriesresonant configuration of the circuit 700 of FIG. 7J, the current levelI₀ approximately equals |I_(C)|:

I ₀ ≈|I _(C)|  (244)

resulting in a voltage across the measurement port 708:

V≈|Z _(11,0) |I ₀ ≈R _(Ls) I _(C)  (245)

and in a drive power level:

P≈VI ₀ ≈R _(Ls) |I _(C)|  (246)

For the parallel resonant configuration of the circuit 700 of FIG. 7J,the level |I_(C)| of the current through the sense electrode 702 isapproximately Q_(p) times higher than the drive current level I₀providing:

I ₀ ≈|I _(C)|(ω_(p) L _(s)−(1/ω_(p) C))G _(p) ≈|I _(C)|ω_(p) L _(p) G_(p) ≈|I _(c)|(1+n _(L))/Qp  (247)

The voltage across the measurement port 708 becomes approximately:

V≈I ₀ /|Y _(11,0) |≈|I _(C)ω_(p) L _(p)  (248)

and the power:

P≈VI ₀≈(R _(Ls) +R _(Lp))|I_(C)|²  (249)

In a further aspect, the differential narrowband extrinsic SNR of theseries resonant configuration of the circuit 700 of FIG. 7J may beexpressed as:

ΔSNR _(ex,s) ≈|ΔZ _(r) ||I _(C) |/V _(sn)≈(|I _(C) |/V _(sn))|ΔZ_(r)′|/(ω_(s) C)  (250)

with |I_(C)| denoting the magnitude of the sense signal current in thesense electrode 702, which approximately equals the current magnitude|I₀|, and V_(sn) the noise voltage capacitively coupled into the senseelectrode 702 as illustrated in FIG. 7G by the series voltage sourceV_(sn). The differential narrowband extrinsic SNR ΔSNR_(ex,s) may alsobe expressed in terms of the normalized reflected impedance |ΔZ_(r)′| asshown on the right side of Equation (250).

Since the sense circuit 701 transforms the voltage drop across ΔZ_(r) toΔV in the same way as it transforms V_(sn) to V_(n), Equation (250) alsoapplies to the parallel resonant configuration as well as to the seriesand parallel resonant configuration of the circuit 700 of FIG. 7J,meaning that:

ΔSNR_(ex,p)=ΔSNR_(ex,s)  (251)

Equations (250) and (251) indicate that the differential narrowbandextrinsic SNR of the series and parallel resonant configuration of thecircuit 700 of FIG. 7J are no function of the Q-factor.

Adding the capacitor 715 with reference to the circuit 710 of FIG. 7Bmay not impact the differential narrowband extrinsic SNR if the senseelectrode current level |I_(C)| in the sense electrode 702 ismaintained. However, the current |I₀| to be delivered by the currentsource 512 may increase by the factor (1+n_(C))/n_(C), hence increasingthe losses in the equivalent series resistances R_(Ls) of the inductor704 by the factor ((1+n_(C))/n_(C))².

In implementations and operations where the noise current I_(0,n) asindicated in FIG. 7J may cause the predominant contribution in V_(n) aspreviously discussed with reference to FIG. 5F, the noise voltage V_(n)for the series resonant configuration is approximately:

V_(n)≈R_(Ls)I_(0,n)  (252)

while the voltage change |ΔV| in presence of an object (e.g., object112) is:

|ΔV|=|I _(C) ||ΔZ _(r) |≈|I ₀ ||ΔZ _(r)|(253)

The differential narrowband intrinsic SNR with respect to the drivecurrent noise I_(0,n) for the series resonant configuration of thecircuit 700 of FIG. 7J may be expressed as:

ΔSNR _(int,s)≈(|I ₀ |/I _(0,n))|ΔZ _(r) |/R _(Ls)=(|I ₀ /I_(0,n))|ΔZ′|  (254)

Equation (254) may also be written in terms of the Q-factor Q_(s) andΔZ_(r)′ as:

ΔSNR _(int,s)≈(|I ₀ |/I _(0,n))Q _(s) |ΔZ _(r)′|  (255)

With the noise current I_(0,n) as the predominant contribution, thenoise voltage V_(n) at parallel resonance becomes approximately:

V _(n) ≈I _(0,n) /|Y _(11,0)|  (256)

Using Equation (256) and applying Equation (70), the differentialnarrowband intrinsic SNR with respect to the noise current I_(0,n) forthe parallel resonant configuration of the circuit 700 of FIG. 7J may beexpressed as:

ΔSNR _(int,p)≈(|I ₀ |/I _(0,n))|ΔZ _(r)|/(R _(Lp) +R _(Ls))=(|I ₀ |/I_(0,n))|ΔY′|(257)

Equation (257) may also be written in terms of the Q-factor Q_(p) andΔZ_(r)′ as:

ΔSNR _(int,p)≈(|I ₀ |/I _(0,n))Q _(p) |ΔZ _(r)′|  (258)

which is a linear function of the Q-factor Q_(p).

Similar considerations may be made for the thermal noise though notrepeated herein for the circuit 700 of FIG. 7J. Equations for theintrinsic narrowband SNR with respect to the thermal noise are providedin the table of FIG. 7N.

Since ΔSNR_(int,s) and ΔSNR_(int,p) are both proportional to themagnitude of the fractional change |ΔZ′| and since adding the capacitor715 reduces the fractional change |ΔZ′| by the factor n_(C)/(1+n_(C)),the differential narrowband intrinsic SNR of the circuit 710 of FIG. 7Bin both the series and parallel resonant configuration may be(1+n_(C))/n_(C) times lower if compared with the circuit 700 of FIG. 7J.For n_(C)=½, it is approximately three times lower.

In another aspect, the broadband extrinsic SNR as defined by Equation(62) with respect to the induced voltage component V_(sW) at thefundamental WPT operating angular frequency ω_(w) is considered.Assuming electric field coupling as the predominant contribution, thedisturbance signal voltage V_(sn) may relate to the WPT coil voltageV_(WPT) as follows:

V _(sn) ≈V _(sW)≈(C _(sW) /C)V _(WPT)  (259)

where C_(sW) denotes the mutual capacitance between the sense electrode702 and the WPT coil (e.g., WPT coil 202 with reference to FIGS. 2 and3). Further assuming:

1/(ω_(W) C)>>ω_(W) L _(s)  (260)

ω_(s)>>ω_(W)  (261)

the disturbance voltage component V_(W) in the voltage V for the seriesresonant configuration of the circuit 700 of FIG. 7J becomesapproximately:

V _(W) ≈V _(sW)ω_(W) Cω _(W) L _(P) ≈V _(Sw)(ω_(W)/ω_(s))² /n_(L)  (262)

The factor (ω_(W)/ω_(s))²/n_(L) may be considered as the attenuation ofthe low frequency induced voltage V_(sW) by the high pass filter effectof the sense circuit 701. Using:

|V|=|I ₀ |R _(s) ≈|I _(C) |R _(Ls)  (263)

the broadband extrinsic SNR with respect to the fundamental WPTfundamental disturbance component V_(sW) for the series resonantconfiguration of the circuit 700 of FIG. 7J may be expressed as:

SNR _(W,s)≈(|I _(C) |/V _(sW))(1/ω_(s) C)(ω_(s)/ω_(W))² n _(L) /Q_(s)  (264)

For a given ratio |I_(C)|/V_(sW) and susceptance ω_(s) C, the broadbandextrinsic SNR for the series resonant configuration of the circuit 700of FIG. 7J is proportional to the inductance ratio n_(L), the square ofthe frequency ratio ω_(s)/ω_(W), but inverse proportional to theQ-factor Q_(s) of the sense circuit 701.

The disturbance voltage component V_(W) in the voltage V for theparallel resonant configuration of the circuit 700 of FIG. 7J becomesapproximately:

V _(n) =V _(W) ≈V _(sW)ω_(W) Cω _(W) L _(p) ≈V _(sW)(ω_(W)/ω_(p))²/(1+n_(L))  (265)

The factor (ω_(W)/ω_(p))²/(1+n_(L)) may be considered as the attenuationof the low frequency induced voltage V_(sW) by the high pass filtereffect of the sense circuit 701. Further, expressing the sense signalvoltage |V| at the angular frequency ω_(p) in terms of the senseelectrode current |I_(C)|:

|V|≈|I _(C)|((ω_(p) C)⁻¹−ω_(p) L _(s))≈|I _(C)|ω_(p) L _(p)  (266)

the broadband extrinsic SNR with respect to the WPT fundamentaldisturbance voltage component V_(sW) for the parallel resonantconfiguration of the circuit 700 of FIG. 7J may be expressed as:

SNR _(W,p)≈(|I _(C) |/V _(sW))(1/ω_(p) C)(ω_(p)/ω_(W))²  (267)

For a given ratio |I_(C)|/V_(sW) and susceptance ω_(p) C, the broadbandextrinsic SNR for the parallel resonant configuration of the circuit 700of FIG. 7J is proportional to the square of the frequency ratioω_(p)/ω_(W), but no function of Q_(p) and n_(L).

For ω_(s)=ω_(p), the following relation may be found between thebroadband extrinsic SNRs of the series and parallel resonantconfigurations of the circuit 700 of FIG. 7G as given by Equations (264)and (267), respectively:

SNR _(W,p) ≈SNR _(W,s) Q _(s) /n _(L)  (268)

where Q_(s) refers to the Q-factor of the series resonant configurationof the circuit 700. From Equation (268), it can be seen that thebroadband extrinsic SNR for the parallel resonant configuration of thecircuit 700 of FIG. 7J may be substantially higher than that of theseries resonant configuration. In an example implementation with L_(p)=L(n_(L)=1), Q_(s)≈Q=30 for the series and parallel resonantconfiguration, the broadband extrinsic SNR of the parallel resonantconfiguration may be 36 dB higher.

It may be appreciated that adding the capacitor 715 with reference toFIG. 7B has virtually no impact on the low frequency disturbance voltageV_(W) as resulting at the measurement port 708 if:

(ω_(W) C _(p))⁻¹>>ω_(W)(L _(s) +L _(p))  (269)

and:

ω_(s)=ω_(p)>>ω_(W)  (270)

holds. The same is true for the sense signal voltage |V| that may beexpressed for the series resonant configuration of the circuit 700 ofFIG. 7J including the capacitor 715 as:

|V|≈I ₀ R _(s) ≈|I _(C)|/(ω_(s) CQ _(Ls))  (271)

if the sense current level |I_(c)| is maintained by adjusting I₀ to:

I ₀ ≈|I _(C)|(1+n _(C))/n _(C)  (272)

Therefore, it may be concluded that the broadband extrinsic SNR given byEquation (264) also applies to the series resonant configuration of thecircuit 710 of FIG. 7B using the parallel capacitor 715.

Equations (201) to (268) may also apply to the circuit 720 of FIG. 7Cwith some minor modifications e.g., by replacing the inductance L_(s) byL_(s)+L_(σ), the series resistance R_(Ls) by R_(Ls)+R_(w), theinductance L_(p) by L_(m), and the series resistance R_(Lp) by R_(m),where L_(σ) denotes the transformer's 726 secondary referred leakageinductance, R_(w) its secondary referred equivalent series resistancewith respect to the conductor losses, L_(m) its secondary referred maininductance, and R_(m) its secondary referred equivalent seriesresistance with respect to the core losses with reference to FIG. 5H.Further, if L_(σ) is a substantial portion of L+L_(σ), the inductanceratio n_(L)=L/L_(p) can be replaced by (L+L_(σ))/L_(m). Though notproven herein, the conclusions drawn from the analysis of the circuit520 of FIG. 5B may also apply to the circuit 720 of FIG. 7C.

The equivalent circuit model 740-1 as illustrated in FIG. 7K comprisesthe sense electrode's 702 capacitance C, the parallel inductor's 744inductance L_(p) and its equivalent parallel conductance G_(Lp), theseries capacitor's 746 capacitance C_(s), an ideal sense signal voltagesource 552, and an ideal current measurement circuit 550. It may beappreciated that in practical implementations, losses in the capacitors(including the sense electrode 702) are generally substantially lowerthan the losses in the parallel inductor 744. Therefore, an equivalentseries resistance of the capacitors 746 and the sense electrode 702 areneglected (not shown) in the equivalent circuit model 740-1 of FIG. 7K.Further, the equivalent circuit model 740-1 includes an admittanceΔY_(r) in parallel to the capacitance C representing the reflectedadmittance of the object 110, 112, or 114 proximate to the senseelectrode 702. (The reflected admittance ΔY_(r) may be regarded as theobject 110, 112, or 114 as illustrated in FIG. 7E abstracted away). Theequivalent circuit model 740-1 also includes a noise current sourceI_(sn) in parallel to the capacitance C representing the noise currentinduced into sense electrode 702 by the electric field as generated whenWPT is active. The noise current I_(sn) may include any low frequencycomponent (e.g., the fundamental of the WPT operating frequency andharmonics thereof) as well as any high frequency component (e.g.,switching noise at the sense frequency). The equivalent circuit model740-1 further indicates the admittance Y₁₁ and the impedance Z₁₁(=1/Y₁₁), the sense signal voltage V₀ with an additive noise voltagecomponent V_(0,n), the sense signal current I with an additive noisecurrent component I_(n), and the measurement port 748 (indicated by theterminal and the dashed line) where the voltage V₀+V_(0,n) is applied,the current I+I_(n) is measured, and where Y₁₁ or Z₁₁ refer to. Becausethe equivalent circuit model 740-1 applies to the circuit 740 of FIG.7E, the reference numeral 740 is used instead in the followingtheoretical analysis.

With the assumption of an identical sense electrode 702 in the circuits700 and 740, the following relations may apply:

ΔY_(r)′=ΔZ_(r)′  (273)

ΔY_(r)≈ΔZ_(r)(ωC)²  (274)

I_(sn)≈V_(sn)ωC  (275)

ΔY_(r)′, ΔZ_(r)′, ΔZ and V_(sn) referring to the normalized reflectedadmittance, the normalized reflected impedance, the reflected impedanceof the object 110 in the sense electrode 702, and the disturbancevoltage V_(sn) with reference to the circuit 700 of FIG. 7J,respectively.

To analyze the series and parallel resonant configuration of the circuit740 of FIG. 7K, the assumptions:

1/ωL _(p) >>G _(Lp)  (276)

|ΔY _(r) |<<G _(Lp)  (277)

are made for a frequency range about the resonant frequency.

In an implementation configured for parallel resonance and with asusceptance:

ωC _(s) >>|Y ₁₁|  (278)

in a frequency range about the resonant frequency, the admittance Y₁₁ atthe measurement port 748 of the circuit 740 of FIG. 7K in presence of anobject (e.g., object 112) may be expressed as:

Y ₁₁ ≈G _(Lp)+(jωL _(p))⁻¹ +jωC+ΔY _(r)  (279)

In absence of a foreign object, a local minimum of |Y_(11,0)(ω)|(parallel resonance) occurs substantially at an angular frequency ωsatisfying:

(jωL _(p))⁻¹ +jωC≈0  (280)

yielding the parallel resonant angular frequency:

ω_(p)≈(L _(p) C)^(−1/2)  (281)

At this frequency, the admittance Y_(11,0) becomes approximately real:

Y _(11,0) ≈Re{Y _(11,0) }=G _(p) ≈G _(Lp)  (282)

with G_(p) denoting the parallel resonant conductance, while theadmittance Y₁₁ in presence of an object (e.g., object 112) isapproximately:

Y ₁₁ ≈G _(p) +ΔY _(r) ≈G _(Lp) +ΔY _(r)  (283)

with ΔY_(r) referring to the reflected admittance as previously definedwith reference to FIG. 7A.

The fractional change ΔY′ for the parallel resonant configuration of thecircuit 740 of FIG. 7K becomes approximately:

ΔY′≈ΔY _(r) /G _(p) ≈×G _(p) ≈ΔY _(r) /G _(Lp)  (284)

Defining the normalized reflected admittance as:

ΔY _(r) ′=ΔY _(r)/(ω_(p) C)  (285)

the Q-factor of the parallel inductor 744:

Q _(Lp)=1/(ω_(p) L _(p) G _(Lp))  (286)

and the Q-factor of the parallel resonant configuration of the sensecircuit 741 of FIG. 7K as:

Q _(p)≈ω_(p) C/G _(p) ≈Q _(Lp)  (287)

the fractional change may also be written in terms of ΔY_(r)′ and Q_(p):

ΔY′=Q_(p)ΔY_(r)′  (288)

According to Equation (288) with (214), the parallel resonantconfiguration of the circuit 740 of FIG. 7K is equivalent to the seriesresonant configuration of the circuit 700 of FIG. 7J with respect to thefractional change.

To analyze the series resonant configuration of the circuit 740 of FIG.7K, the additional assumption:

|ωC _(p)−(ωL _(p))⁻¹ |>>G _(Lp)  (289)

is made for a frequency range about the resonant frequency. Theimpedance Z₁₁ at the measurement port 748 in presence of an object(e.g., object 112) may be expressed as:

Z ₁₁=(jωC _(s))⁻¹+(G _(Lp)+(jωL _(p))⁻¹ +jωC+ΔY _(r))⁻¹  (290)

Using the approximation of Equation (38), the impedance Z₁₁ may beapproximated as:

Z ₁₁≈(jωC _(s))⁻¹+(jωC+(jωL _(p))⁻¹)⁻¹+(G _(Lp) +ΔY _(r))/(ωC−(ωL_(p))⁻¹)²  (291)

In absence of a foreign object, a local minimum of |Z_(11,0)(ω)| (seriesresonance) occurs substantially at an angular frequency ω satisfying:

(jωL _(p))⁻¹ +jω(C+C _(s))≈0  (292)

yielding for the series resonant angular frequency:

ω_(s)≈(L(C+C _(s)))^(−1/2)  (293)

At this frequency, the impedance Z_(11,0) becomes substantially real:

Z _(11,0) ≈Re{Z _(11,0) }=R _(s) =G _(Lp)/(ω_(s) C _(s))²  (294)

with R_(s) denoting the series resonant resistance, while the impedanceZ₁₁ in presence of an object (e.g., object 112) is approximately:

Z ₁₁ ≈R _(s) +ΔZ≈(G _(Lp) +ΔY _(r))/(ω_(s) C _(s))²  (295)

with:

ΔZ≈ΔY _(r)/(ω_(s) C _(s))²  (296)

Further, the Q-factor of the parallel inductor 744 may be defined as:

Q _(Lp)=1/(ω_(s) L _(p) G _(Lp))  (297)

and the capacitance ratio:

n _(C) =C/C _(s)  (298)

Equation (294) at ω_(s) may be expressed as:

Z _(11,0) =R _(s) ≈n _(c)(1+n _(C))/(Q _(Lp)ω_(s) C)  (299)

For n_(C)>>1, the series resonant resistance R_(s) becomesapproximately:

R _(s) ≈n _(C) ²/(Q _(Lp)ω_(s) C)  (300)

According to Equation (299), the impedance Z₁₁ at ω_(s) of the sensecircuit 741 of FIG. 7K can be increased or reduced by adjusting thecapacitance ratio n_(C)=C/C_(s) accordingly, while maintaining seriesresonance substantially at the nominal sense frequency. Therefore, insome implementation the series resonant configuration of the circuit 740of FIG. 7K is employed as an alternative to using a transformer (e.g.,transformer 726 of FIG. 7C) for transforming the impedance Z₁₁ to bewithin a suitable operating impedance range as previously discussed withreference to FIG. 7C.

The fractional change ΔZ′ for the series resonant configuration of thecircuit 740 of FIG. 7K becomes approximately:

ΔZ′=ΔZ/R _(s) ≈ΔY _(r) /G _(Lp)  (301)

According to Equation (301), the impedance change ΔZ is proportional tothe reflected admittance ΔY_(r). Therefore, the angle arg{ΔZ} of themeasured impedance change ΔZ is indicative of the angle arg{ΔY_(r)}. Insome implementations, the accuracy of the measured angle is improved byapplying the angle correction based on the calibration procedure aspreviously described with reference to the circuit 700 of FIG. 7A.

The Q-factor of the series resonant configuration of the sense circuit741 of FIG. 7K may be defined as:

Q _(s)≈ω_(s)(C+C _(s))/G _(Lp)≈1/(ω_(s) L _(p) G _(Lp))=Q _(Lp)  (302)

which equals the Q-factor of the parallel inductor 744. Usingdefinitions above, Equation (302) may also be expressed in terms of theseries resonant resistance R_(s), the electrode 702 capacitance C, andthe capacitance ratio n_(C) as:

Q _(s) 26 n _(C)(1+n _(C))/(R _(s)ω_(s) C)  (303)

Further, the normalized reflected admittance may be defined as:

ΔY _(r) ′=ΔY _(r)/(ω_(s) C)  (304)

The fractional impedance change ΔZ′ may also be written in terms ofQ_(s) and ΔY_(r)′ as:

ΔZ′≈Q _(s) ΔY _(r) ′n _(C)/(1+n _(C))  (305)

In some implementations based on the circuit 740 of FIG. 7K, the voltagelevel V₀ of the voltage source 542 is adjusted to match a specifiedcurrent |I_(C)| in the sense electrode 702. Therefore, in an aspect, therequired voltage level V₀, the resulting current I at the measurementport 748, and the drive power level P may be considered. For theparallel resonant configuration of the circuit 740 of FIG. 7K, thevoltage level V₀ approximately equals the voltage across the senseelectrode 702 providing the relation:

V ₀ ≈|I _(C)|/(ω_(p) C)  (306)

The current I at the measurement port 748 becomes approximately:

I≈|Y _(11,0) |V ₀ ≈G _(Lp) V ₀  (307)

and the drive power level:

P=V ₀ I≈|I _(C)|² G _(Lp)/(ω_(p) C)² =|I _(c)|² Q _(Lp)/(ω_(p) C)  (308)

For the series resonant configuration of the circuit 740 of FIG. 7K, thevoltage |V_(C)| across the sense electrode 702 is approximately:

|V _(C) |≈|I _(C)|/(ω_(s) C)  (309)

yielding for the drive power level:

P≈|V _(C)|² G _(Lp) ≈|I _(c)|² G _(Lp)/(ω_(s) C)² ≈|I _(c) ² Q_(Lp)/(ω_(s) C)  (310)

which equals the drive power level of the parallel resonantconfiguration of the circuit 740 of FIG. 7K.

In a further aspect, it may be meaningful to define the narrowband SNRat the measurement port 748 of the circuit 740 of FIG. 7K as given byEquation (14), where |ΔI| denotes the magnitude of the current change inthe measured current I due to the presence of an object (e.g., object112) and in the additive noise voltage component as indicated in thecircuit 740 of FIG. 7K. More specifically, the current change |ΔI| mayrefer to the r.m.s. current and in to the r.m.s. noise current asmeasured at the nominal sense frequency in the bandwidth B_(m) of thecurrent measurement circuit 550. This noise current in may includecircuit intrinsic and extrinsic noise components as discussed above. TheSNR as given by Equation (14) is referred herein as to the differentialnarrowband SNR.

In another aspect, it may be meaningful to define the broadbandextrinsic SNR at the measurement port 748 of the circuit 740 of FIG. 7Kas given by Equation (147), where |I| denotes the magnitude of the sensesignal current and /_(w) the disturbance current at the fundamental WPToperating frequency, which may be a prominent component in I_(n) whenWPT is active. More specifically, the current |I| may refer to ther.m.s. current and I_(W) to the r.m.s. disturbance current as measuredat the measurement port 748 at the fundamental WPT operating frequency.

Using Equation (14), the differential narrowband extrinsic SNR of theparallel resonant configuration of the circuit 740 of FIG. 7K may beexpressed as:

ΔSNR _(ex,p)≈(|I _(C) |/I _(sn))(|ΔY _(r)|/ω_(p) C)=(|I _(C) |/I_(sn))|ΔY_(r)′|  (311)

with I_(sn) the noise current as illustrated in FIG. 7K.

Since the sense circuit 741 transforms the shunt current through ΔY_(r)to the current change ΔI in the same way as it transforms I_(sn) toI_(n), Equation (311) also applies to the series resonant configuration,meaning that:

ΔSNR _(ex,s) ≈ΔSNR _(ex,p)  (312)

In implementations with the noise voltage V_(0,n) causing thepredominant noise contribution in I_(n) as previously discussed, thenoise current I_(n) for the parallel resonant configuration of thecircuit 740 is approximately:

I _(n) ≈G _(p) V _(0,n)  (313)

while the current change in presence of an object (e.g., object 112) is:

|ΔI|=|V_(L) ||ΔY _(r) |≈|V ₀ ||ΔY _(r)|  (314)

Applying Equations above to Equation (14), the differential narrowbandintrinsic SNR with respect to the noise voltage V_(0,n) for the parallelresonant configuration of the circuit 740 of FIG. 7K may be expressedas:

ΔSNR _(int,p)≈(|V ₀ |/V _(0,n))|ΔY _(r) |/G _(p)  (315)

Equation (315) may also be written as:

ΔSNR _(int,p)≈(|V ₀ |/V _(0,n))Q _(p) |ΔY _(r)′|(316)

Using:

ΔZ′<<1  (317)

the current change magnitude |ΔI| for the series resonant configurationof the circuit 740 of FIG. 7G may be written as:

|ΔI|≈|V ₀ ||ΔZ|/|Z _(11,0)|²  (318)

With the noise voltage V_(0,n) as the predominant noise contribution,the noise current I_(n) at series resonance becomes:

I _(n) =V _(0,n) /|Z _(11,0)|  (319)

The differential narrowband intrinsic SNR with respect to the noisevoltage V_(0,n) for the series resonant configuration of the circuit 540of FIG. 5G may be expressed as:

ΔSNR _(int,s)≈(|V ₀ |/V _(0,n))|ΔZ′|≈(|V ₀ |/V _(0,n))|ΔY_(r) |/G_(Lp)  (320)

Equation (320) may also be written in terms of the Q-factor Q_(s) andthe normalized reflected admittance ΔY_(r)′ as:

ΔSNR _(int,s)≈(|V ₀ |/V _(0,n))Q _(s) |ΔY _(r) ′|n _(C)/(1+n_(C))  (321)

Similar considerations may be made for the thermal noise though notrepeated herein for the circuit 740 of FIG. 7K. An equation for thedifferential narrowband intrinsic SNR with respect to thermal noise isgiven in the table of FIG. 7N.

In a further aspect, the broadband extrinsic SNR as defined by Equation(147) with respect to the induced current component:

I_(sn)=I_(sW)  (322)

at the fundamental WPT operating angular frequency ω_(W) is considered.Further, assuming:

1/(ω_(W) C)>>ω_(W) L _(p)  (323)

the disturbance current component I_(W) in the current I for theparallel resonant configuration of the circuit 740 of FIG. 7K becomesapproximately:

I _(n) =I _(W) ≈I _(sW)ω_(W) L _(p)ω_(W) C _(s) ≈I _(sW) (ω_(W)/ω_(p))²/n _(c)  (324)

The factor (ω_(W)/ω_(p))²/n_(C) may be considered as the attenuation ofthe low frequency induced current I_(sW) by the high pass filter effectof the sense circuit 741. Using:

|I|≈|V _(C) |G _(Lp)  (325)

the broadband extrinsic SNR of the parallel resonant configuration ofthe circuit 740 of FIG. 7K may be expressed as:

SNR _(W,p)≈(|V _(C) |G _(Lp) /I _(sW))(ω_(p)/ω_(W))² n _(C)  (326)

with:

|V _(C) |≈|I _(C)|/(ω_(p) C)  (327)

representing the voltage across the sense electrode 702. Equation (326)may also be written in terms of the Q-factor Q_(p) and the inductanceratio n_(C) as:

SNR _(W,p)≈(|I _(C) |/I _(sW))(ω_(p)/ω_(W))² n _(C) /Q _(p)  (328)

The disturbance current I_(W) in the current I for the series resonantconfiguration of the circuit 740 of FIG. 7K becomes approximately:

I _(W) ≈I _(sW)ω_(W) Lω _(W) C _(s) ≈I _(sW)(ω_(W)/ω_(s))²/(1+n_(C))  (329)

The factor (ω_(W)/ω_(p))²/(1+n_(C)) may be considered as the attenuationof the low frequency induced current I_(sW) by the high pass filtereffect of the sense circuit 741. Further, expressing the sense signalcurrent |I| at the angular frequency ω_(s) in terms of the senseelectrode 702 voltage |V_(C)| is as follows:

|I|≈|V _(C)|(ω_(s) C−(ω_(s) L _(p))⁻¹)≈|V_(C)|ω_(s) C _(s)  (330)

The broadband extrinsic SNR with respect to the WPT fundamentaldisturbance current component I_(sw) for the series resonantconfiguration of the circuit 740 of FIG. 7K may be expressed as:

SNR _(W,s)≈(|V _(C)|ω_(s) C _(s) /I _(sW))(ω_(s)/ω_(W))²(1+n_(C))  (331)

Using the relation:

|V _(C) |≈|I _(C)|/ω_(s) C  (332)

Equation (331) may also be written as:

SNR _(W,s)≈(|I _(C) |//I _(sW))(ω_(s)/ω_(W))²(1+n _(C))/n _(C)  (333)

Based on Equations (328) and (333) and ω_(s)=ω_(p), the followingrelation between the broadband extrinsic SNRs of the parallel and seriesresonant configurations of the circuit 740 of FIG. 7K may be found:

SNR _(W,s) ≈SNR _(W,p) Q _(p)(1+n _(C))/n _(C) ²  (334)

A selection of equations with respect to the resonant frequency, theQ-factor, the impedance/admittance of the sense circuit, the fractionalchange, and the various SNRs for the series and parallel resonantconfigurations of the circuit 700 of FIG. 7J and the circuit 740 of FIG.7K are listed in the table of FIG. 7N. As previously noted, theseequations are valid for the assumptions made with reference to FIGS. 7Jand 7K.

TABLE 3 provides example parameter values as used for a numericalanalysis of the series and parallel resonant configuration of thecircuit 700 of FIG. 7J and the circuit 740 of FIG. 7K. Values for theinduced disturbance voltage V_(sW), the noise voltage V_(sn), and theequivalent respective currents I_(sW) and I_(sn) of the circuit 740 maybe considered typical for the multi-purpose detection circuit 100integrated into a wireless power transfer structure (e.g., wirelesspower transfer structure 200 with reference to FIG. 2). The normalizedreflected impedance of the object 114 as given in TABLE 1 may be typicalfor a human extremity (e.g., a hand) in a distance of 150 mm from asingle-ended sense electrode 702 (e.g., capacitive sense element 109 a)composed of two sections (as illustrated in FIG. 1) each with a formfactor of about 350×45 mm and electrically connected in parallel. Theexample sense current level |I_(C)| may be within a constraint given byan electromagnetic emission limit of an established EMC standard (e.g.,EN 300330). The example SNRs |I₀|I_(0,n) and |V₀|/V_(0,n) may be typicalfor a digital implementation of a sense signal source (e.g., sensesignal current source 512 and sense signal voltage source 552),respectively, as previously described with reference to FIGS. 4 and 5F.

TABLE 3 Circuit 700 of FIG. 7J 740 of FIG. 7K Series Parallel SeriesParallel Configuration resonant resonant resonant resonant Nominal sense3 MHz 3 MHz 3 MHz 3 MHz frequency WPT operating 85 kHz 85 kHz 85 kHz 85kHz frequency Capacitance C of sense 30 pF 30 pF 30 pF 30 pF electrode702 Inductance/capacitance n_(L) = 1 n_(L) = 2 n_(C) = 2.5 n_(C) = 1ratio Q-factor of inductor Q_(Ls) = 30 Q_(Ls) = 30 QL_(p) = 30 Q_(Lp) =30 704/inductor 744 Q-factor of sense Q >> Q_(Ls) Q >> Q_(Ls) Q >>Q_(Lp) Q >> Q_(Lp) electrode 702 Q-factor of inductor Q_(Lp) = 30 Q_(Lp)= 30 Q_(Cs) >> Q_(Lp) Q_(Cs) >> Q_(Lp) 706/capacitor 746 Normalizedreflected |ΔZ_(r)′| = |ΔZ_(r)′| = |ΔY_(r)′| = |ΔY_(r)′| = impedance 100ppm 100 ppm 100 ppm 100 ppm Angle of reflected arg{ΔZ_(r)} = arg{ΔZ_(r)}= arg{ΔY_(r)} = arg{ΔY_(r)} = impedance arg{ΔZ_(r)} 45° 45° 45° 45°Sense electrode 702 5 mA_(rms) 5 mA_(rms) 5 mArms 5 mA_(rms) currentlevel |I_(C)| Extrinsic noise voltage 25 μV_(rms) 25 μV_(rms) 14.1 nArms14.1 nA_(rms) V_(sn)/current I_(sn) (WPT switching noise) SNR of sensesignal |I₀|/I_(0, n) = |I₀|/I_(0, n) = |V₀|/V_(0, n) = |V₀|/V_(0, n) =source 512/552 80 dB 80 dB 80 dB 80 dB Ambient temperature T 350 K 350 K350 K 350 K Equiv. noise bandwidth 200 Hz 200 Hz 200 Hz 200 Hz B_(m) ofmeasurement circuit 510/540 WPT fundamental 150 V_(rms) 150 V_(rms) 2.4mA_(rms) 2.4 mA_(rms) disturbance voltage V_(sW)/current I_(sW)

Numerical results as obtained based on the numerical assumptions ofTABLE 3 using the relevant equations as defined above with reference toFIGS. 7J and 7K are listed in TABLE 6. TABLE 6 additionally includesnumerical results for the angle error arg{ΔZ′} as noted in connectionwith FIG. 7A. Further, it includes the drive current level I₀, the drivepower level P to drive the sense electrode 702 of the sense circuit 701with the sense current |I_(C)| as specified in TABLE 5. Accordingly, itincludes the drive voltage level V₀, the drive power level P to drivethe sense electrode 702 of the sense circuit 741 with the sense currentl/cl as specified in TABLE 5. Some of the results listed in TABLE 6 weredetermined using a circuit analysis tool.

TABLE 4 Circuit 700 of FIG. 7J 740 of FIG. 7K Series Parallel SeriesParallel Configuration resonant resonant resonant resonant Inductance ofinductor L_(s) = 93.8 μH L_(s) = 62.5 μH L_(p) = 67 μH L_(p) = 93.8 μH704/744 Inductance/capacitance of L_(p) = 93.8 μH L_(p) = 31.3 μH C_(s)= 12 pF C_(s) = 30 pF inductor706/ capacitor 746 Q-factor of sensecircuit Q_(s) ≈ 30 Q_(p) ≈ 30 Q_(s) ≈ 30 Q_(p) ≈ 30 501/541 Precisefrequency of 3.0017 MHz 2.9967 MHz 2.9959 MHz 3.0017 MHz minimum|Z_(11.0)|/|Y_(11.0)| Impedance |Z_(11.0)| of sense 58.8 Ω 5.9 kΩ 511 Ω53.1 kΩ circuit 501/541 Fractional change |ΔZ′| 0.30% 0.30% 0.21% 0.30%Impedance angle error ε −0.04°  −1.9°   −1.9°   −0.04°  Drive currentI₀/ ≈5 mA_(rms) ≈0.5 mA_(rms) ≈1.0 V_(rms) ≈8.8 V_(rms) drive voltage V₀Voltage across |Z_(11.0)|/ ≈0.29 V_(rms) ≈3 V_(rms) ≈2 mA_(rms) ≈0.17mA_(rms) current through |Z_(11.0)| Drive power P ≈1.5 mW ≈1.5 mW ≈2 mW≈1.5 mW Differential narrow-band ΔSNR_(ex, s) ≈ ΔSNR_(ex, p) ≈ΔSNR_(ex, s) ≈ ΔSNR_(ex, p) ≈ extrinsic SNR (WPT switching 31 dB 31 dB31 dB 31 dB noise) Differential narrow-band ΔSNR_(int, s) ≈ΔSNR_(int, p) ≈ ΔSNR_(int, s) ≈ ΔSNR_(int, p) ≈ intrinsic SNR (Sensesignal 29.5 dB 29.5 dB 26.6 dB 29.5 dB noise) Differential narrow-bandΔSNR_(int, s) ≈ ΔSNR_(int, p) ≈ ΔSNR_(int, s) ≈ ΔSNR_(int, p) ≈intrinsic SNR (Thermal noise) 95.3 dB 95.4 dB 95.4 dB 95.4 dB Broadbandextrinsic SNR SNR_(W, s) ≈ SNR_(W, p) ≈ SNR_(W, s) ≈ SNR_(W, p) ≈ (WPTfundamental 7.8 dB 37.3 dB 71.2 dB 38.7 dB disturbance)

Based on numerical results listed in TABLE 4, the following conclusionsmay be drawn. The high impedance magnitude |Z_(11,0)| as generallypresented by the parallel resonant configuration of the circuit 700 ofFIG. 7A can be substantially decreased without loss in fractional changeby configuring the sense circuit 701 with an inductance ratio n_(L)>1(e.g., n_(L)=2). Conversely, the low impedance magnitude |Z_(11,0)| asgenerally presented by the series resonant configuration of the circuit740 of FIG. 7E can be increased at the expense of a moderate loss infractional change by configuring the sense circuit 741 with acapacitance ratio n_(C)>1 (e.g., n_(C)=2.5). Further, the results inTABLE 4 show the circuits and configurations equivalent in terms of thedifferential narrowband extrinsic SNR (ΔSNR_(ex)) e.g., with respect toWPT switching noise. Moreover, the numbers for the differentialnarrowband intrinsic SNR with respect to thermal (resistance) noise showthe circuits and configurations equivalent. They also indicate thatthermal noise may have a negligible impact on the overall noise, evenwhen WPT is inactive. The numbers resulting for the broadband extrinsicSNR with respect to the WPT fundamental disturbance voltage (V_(sW))show a substantial difference (>60 dB) between the series resonantconfiguration of the circuit 700 and 740. The parallel resonantconfiguration of the circuit 700 and 740 are almost equivalent and thebroadband extrinsic SNRs of the circuits and configurations are above 6dB, which may be a minimum requirement in a practical implementation.TABLE 4 further shows a negligible angle error |ε| for the seriesresonant configuration of the circuit 700 and the parallel resonantconfiguration of the circuit 740 and an angle error of about 2° for eachof the other configurations. With the exception of the parallel resonantconfiguration of the circuit 740, the current or voltage levels asrequired at the respective measurement port 708 and 748 for driving thesense electrode 702 with the specified sense current level |I_(C)| maybe within suitable ranges of low power electronics for the othercircuits and configurations as theoretically analyzed herein. Thevoltage as required to drive the circuit 740 in the parallel resonantconfiguration may exceed a constraint as given by electronic circuitryand may require transformation e.g., using the transformer 726.

FIG. 7L illustrates a “π”-equivalent circuit model applicable to thecapacitive sense elements used in the 760, 770, and 780 of FIGS. 7G, 7H,and 7I, respectively. The circuit model 762-1 comprises threecapacitances connected in a “π”-topology and related to the capacitancesC₁, C₂, and the mutual capacitance C_(M) as indicated in FIGS. 7G, 7H,and 7I.

FIG. 7M illustrates another equivalent circuit model 762-2 applicable tothe capacitive sense elements used in the circuits 760, 770, and 780 ofFIGS. 7G, 7H, and 7I, respectively. The circuit model 762-2 comprisesthe capacitances C₁ and C₂ in parallel to the respectivevoltage-controlled current sources I_(ind,1) and I_(ind,2) representingthe current induced into the primary and secondary sense electrode,respectively. As with the equivalent circuit models 762-1 of FIG. 7L and562-1 of FIG. 5I, the equivalent circuit model 762-2 of FIG. 7M iselectrically dual to the equivalent circuit model 562-2 of FIG. 5J.

FIG. 7N shows a table of a summary of selected equations with respect tothe resonant frequency, the Q-factor and the impedance or admittance ofthe sense circuit, fractional change, and the various SNRs for theseries and parallel resonant configurations of the circuit 700 of FIG.7J and the circuit 740 of FIG. 7K. As previously noted, these equationsare valid for the assumptions made with reference to FIGS. 7J and 7K.

FIG. 8A illustrates a complex plane 800 or more precisely a portion ofthe complex plane comprising quadrant 1 where the reflected admittanceΔY_(r) of different types (categories) of objects (e.g., object 110,112, 114, or vehicle 330) may occur if proximate to a sense electrode(e.g., sense electrode 702 with reference to FIG. 7E). Morespecifically, FIG. 8A shows shaded areas (angle ranges 802 and 804)where the reflected admittance ΔY_(r) of different types (categories) ofobjects (e.g., object 110, 112, 114) may be measured at a sensefrequency (e.g., in the MHz range). To emphasize the characteristics ofthe different categories of objects, the angle ranges 802 to 804indicated in FIG. 8A may be not drawn to scale and should be consideredqualitative rather than quantitative. There may exist objects (e.g.,110, 112, 114, or vehicle 330) reflecting an admittance ΔY_(r) withangles outside the ranges 802 and 804. Moreover, the actual angle rangesmay also depend on the particular sense frequency, certaincharacteristics of the capacitive sense element (e.g., sense element702), the inductive sensing effect of the capacitive sense element aspreviously discussed with reference to FIG. 1, the position andorientation of an object relative to the capacitive sense element.

The complex plane 800 and the shaded areas (e.g., angle ranges 802 and804) may also apply to the reflected impedance ΔZ_(r) by simplyrelabeling the real and imaginary axis by Re{ΔZ_(r)} and jIm{ΔZ_(r)},respectively (not shown in FIG. 8A).

Further, FIG. 8A illustrates a metallic object 110 represented by a 1€cent coin (object 110 a), two different types of non-living dielectricobjects 112 such as a piece of plastic (object 112 c) and water drops(object 112 d). Moreover, it illustrates a living object 114representing a hand (symbolizing a human extremity). FIG. 8A shows themetallic object 110 a, the dielectric object 112 c, and the livingobject 114 associated with the angle range 802. Further, FIG. 8A showswater drops (object 112 d) associated with the angle range 804.

In some implementations of the multi-purpose detection circuit 100,water dripping from a wet underbody of a vehicle (e.g., vehicle 330)onto the housing of a wireless power transfer structure(e.g., housing328 of wireless power transfer structure 200 of FIG. 3) integrating acapacitive sense element (e.g., capacitive sense element 109 a) maycause a false positive detection. Therefore, it may be desirable todiscriminate water drops (object 112 d) based on the angle arg{ΔY,} toprevent a false positive detection due to the water drops. Thepeculiarities of water drops with respect to the reflected admittanceΔY_(r) are analyzed below with reference to FIGS. 8B, 8C, 8D, 8E, and8F.

The effect of an object (e.g., object 110, 112, 114, or vehicle 330)proximate to a single-ended capacitive sense element (e.g., senseelectrode 702) having a signal terminal 703 may be modeled empiricallyby a ground-related one-port equivalent circuit model 810 illustrated inFIG. 8B . This circuit is based on the “π”-equivalent circuit model ofcapacitive coupling illustrated in FIG. 7L. The equivalent circuit model810 comprises a primary equivalent parallel capacitance C_(∞)-C_(M)referring to the sense electrode 702, a secondary equivalent parallelcapacitance C_(ob)-C_(M) and an equivalent parallel conductance G_(ob),both referring to the object (e.g., object 110, 112, or 114), and amutual capacitance C_(M) representing the capacitive coupling betweenthe sense electrode 702 and the object. More specifically, thecapacitance C_(∞) refers to the capacitance as it may be measured at thesignal terminal 703 in presence of an object (e.g., object 110, 112, or114) with at least one of an infinite capacitance C_(ob) and infiniteconductance G_(ob). The capacitance C_(∞) may also be considered as thesecondary short-circuit capacitance. Likewise, the capacitance C_(ob)refers to the capacitance of the object in presence of the senseelectrode 702 shortened to ground at its signal terminal 703. Thecapacitance C_(ob) may also be considered as the primary short-circuitcapacitance. Further, the capacitances C_(ob) and C_(M) may relate tothe object's size, geometry, position and orientation relative to thesense electrode 702, but also to electrical characteristics of thematerials the object is composed of.

It is assumed that any real heterogenous object (e.g., object 110, 112,or 114) composed of different materials may be substituted by anequivalent homogenous object consisting of a material having a complexrelative permittivity defined as:

ε_(r)=ε_(r) ′−jε _(r)″  (335)

where ε_(r)′ refers to the relative real permittivity and ε_(r)″ to therelative imaginary permittivity related to the electrical losses of thematerial. The relative imaginary permittivity may comprise a dielectricloss coefficient ε_(d,r)″ attributed to bound charge and dipolerelaxation phenomena of the material and another loss coefficientattributed to the material's electrical conductivity σ. The relativeimaginary permittivity may be defined as:

ε_(r)″=ε_(d,r)″+σ/(ωε₀)  (336)

In general, the complex relative permittivity of the equivalent objectwill depend on the position and orientation of the real object relativeto the capacitive sense element (e.g., sense electrode 702). The ratioε_(r)″/ε_(r)′ is commonly known as the loss tangent of a dielectricmaterial:

tan δ=ε_(r)″/(ε_(d,r)″+σ/(ωε₀))  (337)

In the equivalent circuit model 810 of FIG. 8B, the object (e.g., object110, 112, 114, or vehicle 330) is abstracted by an admittanceY_(ob)=G_(ob)+jωC_(ob). It is further assumed that the object admittanceY_(ob) is a function of the complex relative permittivity ε_(r) of theequivalent homogenous object as follows:

Y _(ob) =G _(ob) +jωC _(ob) =JωC _(ob0) f(ε_(r))  (338)

where C_(ob,0) refers to the capacitance of a fictitious (“stealth”)object of a material with ε_(r)′=1 and ε_(r)″=0 (σ=0) that isindistinguishable from air and thus from the absence of a foreignobject. By definition, the function f(1−j0)=1. Further, the functionf(ε_(r)) of the object and maybe of the form:

f(ε_(r))=(ε_(r) +a−1)/a=(χ+a)/a  (339)

where the factor α depends on the geometry, position, and orientation ofthe object (e.g., object 112) and χ denotes the electric susceptibilityof the dielectric material of the equivalent object. For objects withsimilar length in all three dimensions (e.g., a sphere-like shapedobject), the factor a may be of the order of 3. It may be useful todefine an effective complex relative permittivity of the object as:

ε_(r,eff) =f(ε_(r))  (340)

For increasing |ε_(r)|>>α, the effective relative permittivity ε_(r,eff)approaches ε_(r)/a.

Based on the equivalent circuit model 810, the admittance as presentedat the signal terminal 703 of the sense electrode 702 in presence of anobject (e.g., object 110, 112, 114) may be found as:

Y=jωC _(∞)+ω² C _(M) /Y _(ob)  (341)

Defining the capacitive coupling factor kc according to Equation (185)and with respect to the capacitance C_(ob,0) as:

k _(C) ² =C _(M) ²/(C _(∞) C _(ob,0))  (342)

and applying Equation (338) to Equation (341) yields:

Y=jωC _(∞)(1+k _(C) ²/ε_(r,eff))  (343)

Further, defining C as the sense electrode's 702 capacitance in absenceof a foreign object (ε_(r,eff)=1), the limit capacitance C_(∞) forε_(r,eff) →∞ may be expressed as:

C _(∞) =C/(1+k _(C) ²)  (344)

and the admittance in absence of a foreign object (ε_(r,eff)=1) as:

Y₀=jωC  (345)

Applying Equation (344) to Equation (343) and using Equation (345),provides the reflected admittance of the object (e.g., object 110, 112,114) in terms of k_(C) and ε_(r,eff):

ΔY _(r) =Y−Y ₀ =jωC(1−1/ε_(r,eff))k _(C) ²/(1−k _(C) ²)  (346)

In a further aspect, it may be useful to define the limit reflectedadmittance ΔY_(r,∞) for the complex permittivity approaching infinity(ε_(r,eff)→∞)″.

ΔY _(r,∞) =jωCk _(C) ²/(1−k _(C) ²)  (347)

and to normalize the admittance ΔY_(r) to the limit reflected admittanceΔY_(r,∞) as follows:

ΔY _(r) /|ΔY _(r,∞) |=j(1−1/ε_(r,eff))  (248)

FIG. 8C illustrates quadrant 1 of another complex plane 820 of thenormalized reflected admittance ΔY_(r)/ΔY_(r), indicating contour linesof constant ε_(r,eff)′ and constant ε_(r,eff)″. This normalizedadmittance chart shows that increasing at least one of ε_(r,eff)′ andε_(r,eff)″ ends up at ΔY_(r)/|ΔY_(r,∞)|=j. Conversely, decreasingε_(r,eff)′ and ε_(r,eff)″ ends up in the origin ΔY_(r)/|ΔY_(r,∞)|=0.FIG. 8C also indicates a normalized reflected admittanceΔY_(r)/|ΔY_(r,∞)| that is substantially imaginary (e.g., in the anglerange 802 with reference to FIG. 8A) for dielectric objects (e.g.,object 112 or 114) with a high effective permittivity (e.g.,ε_(r,eff)′>10) regardless of ε_(r,d)″ and σ. The same may be true for ametallic object (e.g., object 110) characterized by a high conductivityσ. On the other hand, an object (e.g., object 112) composed of amaterial with a low relative permittivity (e.g., ε_(r,eff)′<3) and amodest loss coefficient (e.g., ε_(r,eff)″<3) may reflect an admittancewith an angle arg{ΔY_(r)}<78° (e.g., in the angle range 804).

In a series of lab experiments, various living and non-living objects(e.g., objects 110, 112, and 114) were tested with respect to thereflected admittance ΔY_(r) at a sense frequency of 3 MHz when broughtinto proximity of a capacitive sense element (e.g., sense electrode 702)integrated into a wireless power transfer structure enclosed by aplastic housing (e.g., housing 328 of wireless power transfer structure200 of FIG. 3). The reflected admittance ΔY_(r) was determined bymeasuring, the change ΔZ of the impedance Z₁₁ at the measurement port ofa sense circuit (e.g., measurement port 708 of sense circuit 701 of FIG.7A) using a properly calibrated impedance measurement circuit (e.g.,circuit 700 of FIG. 7A).

More specifically, living objects (e.g., a human body extremity of anadult, of an infant, a cat, etc.) were tested with respect to theirreflected admittance when brought into proximity of the capacitive senseelement inside the housing. All tested living objects produced animpedance change ΔZ corresponding to a reflected admittance ΔY_(r) closeto the imaginary axis (e.g., in the angle range 802 of FIG. 8A). TABLE 5lists the real and imaginary part of the complex permittivity ε_(r) andthe conductivity a of various human tissue types at 3 MHz. These tissuetypes together may constitute a substantial portion of a human bodyextremity. The numbers in TABLE 5 suggest an equivalent dielectricmaterial with a relative permittivity e.g., ε_(r)′>30 and ε_(r)″>36,which, according to Equation (340) and (339) also assuming α≈3, maycorrespond to an effective relative permittivity e.g., ε_(r,eff)′>11 andε_(r)″>12. According to Equation (348), an object with ε_(r)>11−j12 mayreflect a substantially imaginary admittance ΔY_(r) (e.g., in the anglerange 802 of FIG. 8A) as experimentally observed.

TABLE 5 Complex permittivity Conductivity Tissue type ε_(r)′ ε_(r)″ σ[mS/m] Skin 75 360 60 Subcutaneous fat 40 300 50 Muscle 50 3600 600Blood 1000 8400 1400 Bone 30-85 36-660 6-110

A reflected admittance ΔY_(r) virtually at the imaginary axis (e.g., inthe angle range 802 of FIG. 8A) was also measured for metallic objects(e.g., object 110) made of a material with a conductivity e.g., σ>10MS/m as expected from Equation (347).

Further, tests were performed with nonliving dielectric objects (e.g.,object 112) e.g., a piece of plastic with ε_(r)′<3, a polyethyleneterephthalate (PET) plastic bottle filled with distilled water (σ≈0),filled with tap water (σ≈0.5 mS/m), filled with salt water (σ≈40 mS/m),potting soil, wet foliage, snow, and ice. All test objects produced animpedance change ΔZ corresponding to a reflected admittance ΔY_(r)virtually at the imaginary axis (e.g., in the angle range 802 of FIG.8A). TABLE 6 lists ε_(r)′, ε_(r)″, and the loss tangent tan(δ) of solidand liquid dielectric materials. All listed materials except water (tapwater and sea water) exhibit a loss tangent<0.05. According to Equation(347), an object (e.g., object 112) of any of these materials may causea substantially imaginary reflected admittance (e.g., in the angle range802). Despite of its high loss tangent (e.g., tan (δ)>30), water mayreflect an admittance ΔY_(r) close to the imaginary axis due to its highrelative permittivity ε_(r)′∞78 as expected from Equation (347).

TABLE 6 Complex permittivity Material ε_(r)′ ε_(r)″ tan(δ) ABS 2.4-3.8<0.07 <0.017 Polyethylene 2.3 <0.002 <0.001 Polypropylene 2.3 <0.001<0.0005 Polyurethane 3.3-3.9 <0.2 <0.05 Polyvinyl chloride (PVC) 4-8<0.12 <0.015 Polyoxymethylene (POM) 3.7 <0.02 <0.005 Plexiglas 3 <0.15<0.05 Teflon 2 <0.00004 <0.00002 Nylon 3.4-4   <0.32 <0.08 Glass  3-10<02 <002 Hard rubber 3.2-4   <0.06 <0.015 Silicon rubber 2.5-3.2 <0.015<0.005 Wood 1.2-3   <0.12 <0.04 Oil 2.2-2.8 <0.002 <0.0007 Distilledwater ~78 <0.01 <0.0001 Tap water ~78 ~30 ~0.38 Salt water (3.5%salinity) ~78 ~3000 ~38 Ice (T = 268 K) 3-4 <0.15 <0.04

Tests were also performed with water dripping on the plastic housing ofa wireless power transfer structure (e.g., housing 328 of the wirelesspower transfer structure 200 with reference to FIG. 3) integrating acapacitive sense element (e.g., capacitive sense element 109 a of FIG.3). These tests may be representative for rain or melt water drippingfrom the vehicle's underbody (e.g., vehicle 330) onto a sensitive areaabove the capacitive sense element when the vehicle is parked over thewireless power transfer structure. It was observed that tap waterdroplets before and after their impact on the surface of the housingabove the capacitive sense element reflect an admittance with an anglearg{ΔY_(r)} e.g., in the range from 25°-40° (e.g., in the angle range804). This observation differs from the reflected admittances asmeasured using larger quantities of the same water e.g., contained in aplastic bottle as mentioned above. According to Equation (347), areflected admittance in the above angle range implies an effectiverelative permittivity |ε_(r,eff)| and also a relative permittivity ledthat is substantially lower than that of tap water as given in TABLE 6.It appears that the effective permittivity of water reduces as thevolume-to-surface area ratio decreases. This phenomenon may relate tothe surface tension effect of liquids caused by intermolecular forces(e.g., Van der Waals forces), which becomes prevalent at smallvolume-to-surface area ratios (e.g., <5 mm). The volume-to-surface arearatio of a body has the dimension of a length unit (e.g., mm) andlinearly grows with the scale factor of the body.

Specific lab experiments were carried out to further investigate thisphenomenon using a test set up with water contained in a plastic hosedisposed proximate to a capacitive sense element (e.g., capacitive senseelement 109 a). The plastic hose was connected to a water reservoirallowing the water level in the hose to be accurately adjusted. Morespecifically, in a first experiment, the reflected admittance ΔY_(r) wasmeasured for a change of water level by 60 mm in a first plastic hosewith a diameter of 2 mm. In a second experiment, it was measured for achange of water level by 15 mm in a second plastic hose having adiameter of 4 mm. In both experiments, changing the water level by 60 mmand 15 mm, respectively, may be considered equivalent to bringing acylindrically shaped sample of water (e.g., object 112) with a volume of188.5 mm³ into proximity of the capacitive sense element, however with adifferent volume-to-surface area ratio. The volume-to-surface area ratioof the 2×60 mm water sample used in the first experiments amounts to0.49 mm vs. 0.88 mm in the second experiment. For comparison, aspherical water droplet of 4 mm diameter provides a volume-to-surfacearea of 0.66 mm. Further, to investigate the impact of the water'sconductivity a, the first and second experiments were performed withcommercially off-the-shelf distilled water, tap water with about 0.03%of calcium and magnesium ions, and with water of different salinityusing a NaCl solution. Starting with distilled water, the NaClconcentration was successively increased (doubled) in a series ofmeasurements.

FIG. 8D displays the normalized reflected admittances ΔY_(r)/|ΔY_(r,∞)|as determined for the different NaCl concentrations in the complex plane820 also showing the contour lines of constant ε_(r,eff)′ and constantε_(r,eff)″ with reference to FIG. 8C. The limit reflected admittanceΔY_(r,∞) was determined by extrapolating the series of measured datapoints towards an infinite conductivity σ. Further, it illustratesobjects 112, which are the cylindrically shaped water samples used inthe first and second experiment as described above. The circular andrectangular marks refer to the normalized reflected admittance asdetermined for the 2×60 mm and for the 4×15 mm water sample,respectively, using distilled water, tap water, and the NaCl solution atconcentrations as indicated in percentage (%). The table on the rightalso indicates the mass percentage of NaCl dissolved in water, thecorresponding molarity (in moles per liter), and the conductivity σ inmS/m as predicted by theory. The data points displayed in FIG. 8Dsuggest that the relative permittivity ε_(r)′ of water with a low NaClconcentration (e.g., <0.1%) substantially reduces as thevolume-to-surface area ratio decreases (e.g., below 1 mm), but steadilyincreases as the NaCl concentration increases.

To illustrate the variation of the angle arg{ΔY_(r)} over the range oftested ion concentrations in the 2×60 mm water sample, FIG. 8E displaysthe unity reflected admittance ΔY_(r)/|ΔY_(r)| (the reflected admittancenormalized to its magnitude |ΔY_(r)|) in a complex plane 840. FIG. 8Eshows the angle of the reflected admittance ΔY_(r) extremely sensitiveon the ion concentration. A very low NaCl concentration (e.g., <0.01%)is enough to produce an angle that substantially differs from 90°. Asapparent from FIG. 8E, even the very low ion concentration in distilledwater may cause a measurable deviation from 90°. Further, contemplatingFIG. 8E, the angle initially decreases as the NaCl concentrationincreases reaching a minimum of 28.6° at about 0.04% NaCl. As theconcentration is further increased the angle arg{ΔY_(r)} turns aroundand reaches 86.5° at 20.5%. Coincidentally, the tap water samplereflected an admittance with an angle close to the minimum angle.Further, tests performed with water collected from a vehicle's underbodyprovide evidence that rain or melt water on the road splashing to thevehicle's underbody already contains enough dissolved minerals toproduce an angle substantially different from 90° (e.g., <60°).

Likewise, FIG. 8F displays the unity reflected admittanceΔY_(r)/|ΔY_(r)| in the complex plane 840 for the 4×15 mm water sampleshowing an angle variation ratio of 86.5°/57.8°≈1.5 versus78.7°/28.6°≈2.75 as obtained from the first experiment with the 2×60 mmwater sample. The ratio 2.75/1.5≈1.84 resembles the ratio 0.88/0.49≈1.8referring to the volume-to-area ratio of the respective water samplesused in these lab experiments. The available experimental data may notsuffice to derive a law between angle variation ratio and volume-to-arearatio though. However, it suggests that the angle variation ratioreduces as the volume-to-area ratio of the water sample increases.

In a further aspect, the effective electric susceptibility defined as:

χ_(e,eff)−ε_(r,eff)′−1  (349)

and the effective conductivity σ_(eff) of the 2×60 mm and 4×15 mm watersamples are analyzed based on the measured reflected admittances andEquations (337), (335), and (336) assuming that the loss coefficientε_(d,r)″ attributed to bound charge and dipole relaxation is negligibleat the sense frequency of 3 MHz. FIG. 8G displays the ratiosχ_(e,eff)/χ_(e) vs. the ratio σ_(eff)/σ in a log-log diagram 860, whereχ_(e) and σ denote the electric susceptibility and the conductivity,respectively, of water as predicted by theory for the NaClconcentrations as indicated in FIG. 8G, for 3 MHz and for a watertemperature of 25° C. The diagram 860 of FIG. 8G reveals that the ratioχ_(e,eff)/χ_(e) increases with increasing NaCl concentration while theratio σ_(eff)/σ decreases. However, both ratios converge to ˜2.4×10⁻⁴ atlow NaCl concentrations (e.g., <0.02%) for the 2×60 mm water sample andto ˜2×10⁻³ for the 4×15 mm sample.

Finally, the diagram 880 of FIG. 8H displays the relation between theeffective complex permittivity ε_(r,eff)=ε_(r,eff)′−jε_(r,eff)″ and theNaCl concentration for the 2×60 mm and the 4×15 mm water sample. Thediagram 880 shows the real part ε_(r,eff)′ of the effective relativepermittivity of the 2×60 mm water sample close to one at ˜0% NaCl(distilled water) and increasing as the NaCl concentration increasesreaching a value of ˜1.8 at 20.5% NaCl, while the absolute value of theimaginary part (representing the electrical loss) ε_(r,eff)″ variesbetween ˜0 and ˜4.8. For the 4×15 mm water sample, the effectiverelative permittivity ε_(r,eff)″ starts at ˜1.2 for ˜0% NaCl (distilledwater) and reaches ˜3.1 at 20.5% NaCl, while ε_(r,eff)″ varies between˜0 and ˜10.5.

The outcomes of the lab experiments as described above indicate thepotential to discriminate between rain or melt water (e.g., object 112)dripping from the vehicle's (e.g., vehicle 330) underbody and livingobjects (e.g., living object 114) based on the angle arg{ΔY_(r)}Therefore, in some implementations of the multi-purpose object detectioncircuit 100 e.g., based on the circuit 700 of FIG. 7A, detections causedby an impedance change ΔZ with an angle arg{ΔZ} substantially deviatingfrom 90° (e.g., in the angel range 804) are suppressed.

Though described above for measuring a change in an admittance orimpedance, discriminating rain or melt water (e.g., object 112) drippingfrom the vehicle's (e.g., vehicle 330) underbody may also beaccomplished based on other electrical characteristics as they may bemeasured in some implementations of the multi-purpose object detectioncircuit 100 and as mentioned in any of the US patent applications hereinincorporated by reference. As observed in the reflected admittanceΔY_(r), rain or melt water (e.g., object 112) dripping from thevehicle's (e.g., vehicle 330) underbody may also cause a change in anelectrical characteristic different from a change produced by otherobjects (e.g., object 110 and 114).

FIG. 9A is a circuit diagram of a circuit 900 illustrating an exampleimplementation of a portion of a multi-purpose detection circuit 100.The circuit 900 of FIG. 9A illustrates an analog front-end circuitportion of the multi-purpose detection circuit 100 with reference toFIGS. 1 and 4. FIG. 9A excludes various other signal generation,processing, control, and evaluation circuits (e.g., as shown in FIG. 4)that may be needed in some implementations of a multi-purpose detectioncircuit 100. The circuit 900 implements inductive and capacitive sensingmeasuring an impedance based on the current source voltage measurementapproach as previously described in connection with FIGS. 5A and 7A,respectively.

The circuit 900 may be subdivided into a driver circuit 402, a pluralityof inductive sense circuits 106, a plurality of capacitive sensecircuits 108, and a measurement amplifier circuit 404 with reference tothe generic block diagram of FIG. 4. The driver circuit 402 and themeasurement amplifier circuit 404 constitute a portion of themeasurement circuit 104 with reference to FIGS. 1 and 4. The pluralityof inductive sense circuits 106 includes sense circuits 106 a, 106 b, .. . , 106 n (106 n not shown in FIG. 9A for purposes of illustration).The plurality of capacitive sense circuits 108 includes sense circuits108 a, 108 b, . . . , 108 n, (108 a and 108 b not shown in FIG. 9A forpurposes of illustration). The dots indicated in FIG. 9A shall indicatethat the number of inductive sense circuits 106 and/or the number ofcapacitive sense circuits 108 may be greater than three as previouslynoted with reference to FIG. 1.

In the example implementation shown in FIG. 9A, each of the plurality ofinductive sense circuits 106 have an identical circuit topology.Likewise, each of the plurality of capacitive sense circuits 108 have anidentical circuit topology. Therefore, descriptions given below for theinductive sense circuit 106 a also apply to the other inductive sensecircuits (e.g., 106 b) and descriptions given below for the capacitivesense circuit 108 n also apply to the other capacitive sense circuits(e.g., 108 a).

Each of the plurality of inductive sense circuits 106 provides a firstmeasurement port 936 (indicated in FIG. 9A by a terminal) for drivingthe inductive sense circuit (e.g., sense circuit 106) with an electricalcurrent I₁ (as indicated in FIG. 9A) and a second measurement port 937(indicated in FIG. 9A by a terminal) for measuring an electrical voltageV₂ (as indicated in FIG. 9A) e.g., in response to the current I₁.Therefore, the sense circuits 106 may be considered as two-portcircuits. Likewise, each of the plurality of capacitive sense circuits108 provides a first measurement port 938 (indicated in FIG. 9A by aterminal) for driving the capacitive sense circuit (e.g., sense circuit108 n) with the current h (as indicated in FIG. 9A) and a secondmeasurement port 939 (indicated in FIG. 9A by a terminal) for measuringthe voltage V₂ (as indicated in FIG. 9A) e.g., in response to thecurrent I₁. Therefore, the sense circuits 108 may be considered astwo-port circuits.

The driver circuit 402 includes an input multiplexer circuit 910 toselectively (e.g., sequentially) drive each of the plurality of sensecircuits 106 and 108 with the current I₁. Likewise, the measurementamplifier circuit 404 includes an output multiplexer circuit 940configured to selectively (e.g., sequentially) measure the voltage V₂ ineach of the plurality of sense circuits 106 and 108. More specifically,but not indicated in FIG. 9A for purposes of illustration, the current hdriving the sense circuit 106 a may be denoted by I_(1a), the current I₁driving the sense circuit 106 b may be denoted as I_(1b), etc. Likewise,the voltages V₂ in the sense circuits 106 a and 106 b may be denoted byV_(2a) and V_(2b), respectively.

The circuit 900 may be configured and operated in a mode to selectively(e.g., sequentially) measure an intra-sense circuit transimpedance Z₂₁e.g., between the measurement ports 936 and 937 of each of the pluralityof the sense circuits 106. This intra-sense circuit transimpedance Z₂₁and may be defined for the sense circuit 106 a as:

Z _(2a1a) ≈V _(2a) /I _(1a)  (350)

For certain configurations of the sense circuits 106 and 108 (examplegiven below), the two-port transimpedance Z₂₁ substantially equals theone-port impedance Z₁₁ as it may be measured at the first measurementport (e.g., measurement port 936) with the second measurement port(e.g., measurement port 937) open-circuited.

However, the circuit 900 may also be configured and operated in a modeto selectively (e.g., sequentially) measure an inter-sense circuittransimpedance Z₂₁ e.g., between each of a plurality of pairs of sensecircuits associated with neighboring sense elements (e.g., inductivesense element 107 a and 107 b) providing sufficient cross-coupling. Theinter-sense circuit transimpedance Z₂₁ as measured between themeasurement port 936 of sense circuit 106 a and the measurement port 937of sense circuit 106 b may be defined as:

Z _(2a1b) ≈V _(2b) /I _(1a)  (351)

In some implementations or operations, inter-sense circuittransimpedance Z₂₁ measurements are performed between pairs of inductivesense circuits (e.g., inductive sense circuits 106 a and 106 b) andbetween pairs of capacitive sense circuits (e.g., capacitive sensecircuits 108 a and 108 b). For simplicity, the intra-sense circuittransimpedance Z₂₁ herein is often referred to as the impedance Z₁₁ andthe inter-sense circuit transimpedance Z₂₁ as the transimpedance Z₂₁.However, in a strict sense, both Z₁₁ and Z₂₁ may represent atransimpedance.

In an aspect, an object (e.g., object 110) proximate to at least one ofthe neighboring sense elements (e.g., 107 a and 107 b) may change boththe impedance Z₁₁ and the transimpedance Z₂₁. Therefore, additionallymeasuring Z₂₁ may improve the detection reliability of the multipurposedetection circuit 100. Example implementations and operations of objectdetection circuits configured to measure both the impedance Z₁₁ and thetransimpedance Z₁₂ are described in U.S. patent application Ser. No.16/358,534, titled Foreign Object Detection Circuit Using MutualImpedance Sensing, the entire contents of which are hereby incorporatedby reference.

The inductive sense circuit 106 a includes an inductive sense element107 a including a sense coil (e.g., sense coil 502 of FIG. 5A) with aninductance L, a first capacitor (e.g., capacitor 504 of FIG. 5A) withcapacitance C_(s), an inductor (e.g., inductor 506 of FIG. 5A) with aninductance L_(p), a second capacitor 928 with a capacitance C_(b1), anda third capacitor 929 with a capacitance C_(b1). The first capacitor 504is electrically connected in series to the inductive sense element 107 athat also connects to ground. The inductor 506 is electrically connectedin parallel to the series circuit of capacitor 504 and inductive senseelement 107 a. The second capacitor 928 capacitively couples the seriescircuit capacitor 504 and inductive sense element 107 a to themeasurement port 936, while the third capacitor 929 capacitively couplesto the measurement port 937.

In an example implementation or configuration of the circuit 900 of FIG.9A, each of the plurality of inductive sense circuits 106 is configured(tuned) to provide a minimum of the impedance |Z₁₁| (series resonance)substantially at the sense frequency as previously discussed withreference to the circuit 500 of FIG. 5A.

In some implementations, at least the capacitor 504 is of a type with alow temperature coefficient providing high thermal stability (e.g., aNP0-type capacitor) reducing thermal drift of an electricalcharacteristic (e.g., an impedance Z₁₁) as measured at each of theplurality of inductive sense circuits 106 a, 106 b, . . . , 106 n. Inother implementations, the capacitor 504 is a temperature compensationcapacitor configured to compensate at least a portion of a temperaturedrift of the inductive sense element 107 a. In a further aspect, theinductor 506 may use a ferrite core or may be an air coil e.g., forpurposes of a higher linearity.

In yet further implementations of the circuit 900 using a printedcircuit board (PCB), the plurality of inductors 506 is arranged toreduce a magnetic field coupling between neighboring inductors 506,e.g., by an alternating orientation. In yet other implementations, theinductor 506 is electromagnetically shielded to reduce at least one of amagnetic field coupling between neighboring inductors 506 and adisturbance voltage induced into the inductor 506 e.g., by the magneticfield as generated by the wireless power transfer structure (e.g.,wireless power transfer structure 200 with reference to FIGS. 2 and 3).

Moreover, as previously mentioned in connection with the circuit 500 ofFIG. 5A, the first capacitor 504 together with the parallel inductor 506form a 2^(nd) order high pass filter to attenuate a low frequencydisturbance component (e.g., at the WPT frequency) in the voltage V aspreviously discussed with reference to FIG. 5A. The capacitor 504together with the parallel inductor 506 may be configured to attenuatethis low frequency disturbance component to a level e.g., significantlybelow the level of the voltage V₂ in response to the respective currentI₁ at the sense frequency. Therefore, this high pass filter maysubstantially reduce dynamic range requirements in the measurementamplifier circuit 404 and in a further processing (e.g., in the signalprocessing circuit 408 with reference to FIG. 4). It may also reduce anycross-modulation effects between any low frequency signals at the WPToperating frequency (fundamental and harmonics thereof) and the sensesignal at the sense frequency. Cross-modulation may be produced e.g., bynon-linear distortion effects in the measurement amplifier circuit 404.At the sense frequency, this high pass filter may exert a minor impacton the voltage V₂ and thus on the measured impedance Z₁₁ and which maybe corrected in a further processing (e.g., in the signal processingcircuit 408 of FIG. 4). Any phase shift caused by this high pass filtermay be determined e.g., by performing a calibration as previouslydiscussed with reference to FIG. 5A.

The second capacitor 928 may be needed in some implementations to blockany DC flow at the output of the driver circuit 402. In an aspect, thecapacitor 928 may also help to attenuate any residual low frequencyvoltage component (e.g., at the WPT operating frequency) at the outputof the driver circuit 402. Moreover, in some implementations, it mayalso be used to compensate or partially compensate for the effect of thereactance of parallel inductor 506 in the measured impedance (e.g., Z₁₁)and hence to reduce an error in the measured angle arg{ΔZ} as previouslydiscussed with respect to FIG. 5A. Likewise, the third capacitor 929 maybe needed in some implementations to block any DC flow at the input ofthe measurement amplifier circuit 404. In some aspect, the capacitor 929may also help to attenuate any residual low frequency current component(e.g., at the WPT operating frequency) at the input of the measurementamplifier circuit 404. Moreover, in some implementations, it may alsohelp to compensate or partially compensate for the effect of thereactance of the parallel inductor 506 in the measured impedance Z₁₁ asdiscussed with reference to FIG. 5A. In implementations orconfigurations using a capacitor 929 providing a reactance substantiallysmaller than an input impedance of the measurement amplifier circuit404, the intra-sense circuit transimpedance Z₂₁ substantially equals theimpedance Z₁₁ as discussed above.

The impedance |Z₁₁| of the inductive sense circuit 106 a at seriesresonance is assumed in the suitable measuring range of the measurementcircuit (e.g., measurement circuit 104 with reference to FIG. 4).Therefore, the inductive sense circuits 106 as illustrated in FIG. 9A donot include a transformer (e.g., transformer 526 of FIG. 5B).

The capacitive sense circuit 108 n includes a capacitive sense element109 n including a sense electrode (e.g., sense electrode 702 of FIG. 7Cillustrating a single-ended sense electrode) having a capacitance C, aseries inductor 724 (e.g., series inductor 724 of FIG. 7C) having aninductance L_(s), and a transformer 726 (e.g., transformer 726 of FIG.7C) providing a primary and secondary port, a secondary-referred maininductance L_(m) and a voltage transformation ratio 1: n_(VT). Further,it includes a first capacitor 930 with a capacitance C_(b3), and asecond capacitor 931 with a capacitance C_(b4) e.g., for purposes aspreviously discussed with reference to the inductive sense circuits 106of FIG. 9A. The inductor 724 is electrically connected in series to thecapacitive sense element 109 n. The series circuit of inductor 724 andcapacitive sense element 109 n is electrically connected to thetransformer's 726 secondary port that also connects to ground. Thetransformer's 726 primary port is capacitively coupled to themeasurement ports 938 and 939 via capacitors 930 and 931, respectively,and also electrically connects to ground.

In some implementations, the transformer 726 comprises a primary windingand a galvanically isolated secondary winding, both windings wound on acommon ferrite core as indicated in FIG. 10. An example transformer 726configured for a nominal sense frequency in the MHz-range uses atwo-hole ferrite core.

In the example implementation or configuration of the circuit 900 ofFIG. 9A, each of the plurality of capacitive sense circuits 108 isconfigured (tuned) to provide a minimum of the impedance magnitude |Z₁₁|(series resonance) substantially at the sense frequency as previouslydiscussed with reference to the circuit 720 of FIG. 7C.

In some implementations, at least the inductor 704 may be of a type witha low temperature coefficient providing higher thermal stabilityreducing thermal drift of the impedance Z₁₁ as measured at each of theplurality of the capacitive sense circuits 108. In otherimplementations, the inductor 704 is a temperature compensation inductorconfigured to compensate at least a portion of a temperature drift ofthe capacitive sense element 109 n. In further implementations (notshown herein), a supplementary temperature compensation capacitor iselectrically connected in parallel to the capacitive sense element 109 n(e.g., as illustrated by capacitor 715 in FIG. 7B) configured tocompensate at least a portion of the total temperature drift of theinductor 704 and the capacitive sense element 109 n. Further, theinductor 724 may uses a ferrite core or is an air coil providing higherlinearity.

In another aspect and in some implementations of the circuit 900 builton a printed circuit board, the plurality of inductors 724 is arrangedto reduce a magnetic field coupling between neighboring inductors 724and a neighboring inductor 506 of an inductive sense circuit (e.g.,inductive sense circuit 106 a), e.g., by an alternating orientation. Inyet other implementations, at least one of the inductor 724 and thetransformer 726 is electromagnetically shielded to reduce at least oneof a magnetic field coupling between neighboring inductors 724, 506, anda disturbance voltage induced e.g., by the magnetic field as generatedby the wireless power transfer structure (e.g., wireless power transferstructure 200 with reference to FIGS. 2 and 3).

In further implementations, at least a portion of the requiredinductance L_(s) is realized by a leakage inductance of the transformer726 as previously mentioned with reference to the circuit 720 of FIG.7C.

As also mentioned in connection with the circuit 720 of FIG. 7C, thesecondary-referred main inductance L_(m) of the transformer 726 togetherwith capacitance C of the capacitive sense element 109 n form a 2^(nd)order high pass filter to attenuate a low frequency disturbancecomponent in the voltage V₂ as previously discussed with reference toFIGS. 7A, 7C, and the inductive sense circuits 106 of FIG. 9A. Thetransformer 726 may be configured to attenuate this low frequencydisturbance component to a level e.g., significantly below the level ofthe voltage V₂ in response to the respective current I₁ at the sensefrequency. At the sense frequency, this high pass filter may exert aminor impact on the voltage V₂ and thus on the measurement of theimpedance Z₁₁ and which may be corrected in a further processing (e.g.,in the signal processing circuit 408 of FIG. 4). Any phase shift causedby this high pass filter may be determined e.g., by performing acalibration as previously discussed with reference to FIG. 7A.

In a further aspect, the transformer 726 may be employed to transformthe impedance Z₁₁ of the sense circuit 108 n as presented at the seriesresonance into the suitable measuring range of the measurement circuit(e.g., measurement circuit 104 of FIG. 4) by adjusting thetransformation ratio 1:n_(VT) accordingly.

The driver circuit 402 includes a driver amplifier circuit 902, an inputmultiplexer circuit 910 illustrated in FIG. 9A as a plurality ofswitches 911 a, 911 b, . . . , 911 n, and a plurality of seriesresistors 914 and 915. The series resistors 914 connected to theinductive sense circuits 106 have a resistance R_(ser1), while theseries resistors 915 connected to the capacitive sense circuits have aresistance R_(ser3) that generally differs from R_(ser1). Each of theoutputs of the input multiplexer circuit 910 connects to the respectivesense circuit of the plurality of sense circuits 106 and 108 via therespective series resistors 914 and 915. The driver circuit 402 isconfigured to operate as a current source (e.g., current source 512 asdescribed in connection with FIG. 5A) and to selectively (e.g.,sequentially) apply a drive current signal I₁ at the sense frequency toeach of the plurality of inductive sense circuits 106 and to each of theplurality of the capacitive sense circuits 108. The drive current signalI₁ (e.g., a sinusoidal signal) is based on a driver input signal whichmay be an output of the signal generator circuit 406 with reference toFIG. 4. The driver amplifier circuit 902 as illustrated in FIG. 9A byexample includes an amplifier 904 and external resistance circuitrycomprising a first (feedback) resistor 906 and a second resistor 908 foradjusting a gain. In some implementations, the amplifier is at least oneof a low noise operational amplifier and an operational amplifierproviding high linearity. The driver amplifier circuit 902 is configuredto receive the driver input signal and to provide a corresponding outputwith an accurate and stable voltage (a voltage source output). Aspreviously mentioned, in some implementations, a DC voltage may bepresent at any of the plurality of outputs of the driver circuit 402caused e.g., by a DC offset in the amplifier's 904 output voltage or bycertain types of analog switches (e.g., switch 911 a) of the inputmultiplexer circuit 910.

The driver amplifier circuit 902 together with series resistors (e.g.,series resistor 914) mimic a current source characteristic at each ofthe plurality of outputs of the driver circuit 402. It may beappreciated that the series resistor (e.g., series resistor 914) with aresistance (e.g., R_(ser1)) substantially larger (e.g., 10 times larger)than the impedance magnitude |Z₁₁| at series resonance e.g., of thesense circuit 106 may transform the voltage source output of the driveramplifier circuit 902 into a current source output meeting therequirements of a current source 512 as previously defined in connectionwith FIG. 5A. Increasing the series resistance (e.g., R_(ser1)) mayimprove the current source characteristic but results in a lower levelof drive current I₀. The drive current level may impact a SNR aspreviously defined with reference to FIG. 5F. Therefore, in someimplementations, the series resistances R_(ser1) and R_(ser3) mayrepresent a trade-off between a current source characteristic and a SNR.

In an alternative configuration, the current source characteristic isrealized using a resistor (e.g., series resistor 914) with a lowerresistance (e.g., R_(ser1)) instead using the DC block capacitor (e.g.,capacitor 928) with a higher reactance, together providing an impedancesubstantially larger (e.g., 10 times larger) than the impedance aspresented at the primary port of the transformer 726 at seriesresonance. In another implementation variant (not shown herein), theresistor (e.g., series resistor 914) is omitted entirely and the highimpedance is realized by the DC block capacitor (e.g., capacitor 928).In a further implementation variant (not shown herein), the high seriesimpedance as required to mimic a current source characteristic isrealized at least in part by using at least one of an inductor (notshown in FIG. 9A) and a leakage inductance of a transformer.

In yet another implementation variant, the current source characteristicis realized using a driver amplifier circuit 902 configured as aregulated current source. An example current source circuit using anoperational amplifier is illustrated in FIG. 9B. However, the currentsource characteristic as apparent at each of the plurality of outputs ofthe of the driver circuit 402 may be significantly impaired by theparasitic capacitances of the switch (e.g., switch 911 a) as previouslydescribed. Therefore, this implementation variant (not shown herein),may incorporate the input multiplexer circuit 910 into the driveramplifier circuit 902 also employing an additional (third) multiplexercircuit in a feedback path. This implementation variant may provide aregulated (stable) current source characteristic at each of theplurality of outputs of the driver circuit 402 substantially eliminatingthe effect of the switch' parasitic capacitances.

In another implementation variant (not shown herein), the driveramplifier circuit 902 additionally includes an output transformer e.g.,for purposes of transforming an output voltage. As opposed to loweringthe resistance (e.g., R_(ser1)), the use of an output transformer mayallow the drive current level I₁ and thus the sense element currentlevels I_(L) and I_(C) to be increased without compromising the currentsource characteristic of the driver circuit 402

The input multiplexer circuit 910 includes a plurality of switches 911a, 911 b, . . . , 911 n and is configured to selectively connect each ofthe plurality of inductive sense circuits 106 and each of the pluralityof capacitive sense circuits 108 via the respective series resistor 914and 915 to the driver circuit 402 to selectively (e.g., sequentially)drive each of the plurality of inductive sense circuits 106 and each ofthe plurality of capacitive sense circuits 108 with the current I₁ atthe sense frequency. Therefore, each of the plurality of switches 911 a,911 b, . . . , 911 n is electrically connected to the driver amplifiercircuit's 902 output that is also referred to as the common input node.The input multiplexer circuit 910 is further configured to receive aninput MUX control signal from a control circuit (e.g., from the controland evaluation circuit 102 of FIG. 4) that controls the inputmultiplexer circuit 910.

Each of the plurality of switches 911 a, 911 b, . . . , 911 n may be oneof a semiconductor analog switch (e.g., a single field effect transistor(FET) switch, a complementary FET switch composed of a p-channel and an-channel type FET), a micro-mechanical systems (MEMS) switch or anyother type of switch providing a sufficiently low series resistance,when the switch is closed (closed-state) and a sufficiently high currentsignal attenuation, when the switch is open (open-state). An exampleimplementation of an analog switch (e.g., switch 911 a) based on asingle FET is illustrated in FIG. 9D.

The switches (e.g., switch 911 a) may be characterized by a closed-stateseries resistance, an equivalent open-state series capacitance, and anequivalent parallel capacitance at each side of the switch. It may beappreciated that the closed-state resistance of the switch (e.g., switch911 a) is non-critical as it merges with the resistance (e.g., R_(ser1))of the series resistor (e.g., series resistor 914). It may also beappreciated that the total capacitive load produced by the plurality ofparallel capacitances at the common input node may be non-critical sinceit is in parallel to the voltage source output of the driver amplifiercircuit 902.

The switch (e.g., switch 911 a) of an example input multiplexer circuit910 may use complementary FET switches with a closed-state resistance of5 Ω, an equivalent open-state series capacitance of 3 pF (correspondingto a series reactance of 17.71 kΩ at a sense frequency of 3 MHz), and anequivalent parallel capacitance of 12 pF on each side of the switch.

In a further aspect, the closed-state resistance of a semiconductoranalog switch (e.g., switch 911 a) may be subject of a temperature driftthat may impact the temperature stability of the driver circuit 402.Therefore, in some implementations, the impact of the input multiplexercircuit 910 switch (e.g., switch 911 a) is reduced by using a resistor(e.g., series resistor 914) whose resistance (e.g., R_(ser1)) issubstantially larger than the closed-state resistance of the switch.Therefore, in some implementations, the series resistances R_(ser1) andR_(ser3) may also represent a trade-off between a temperature stabilityand a SNR as discussed above.

In an implementation variant of the circuit 900 (not shown herein), theorder of the series resistor (e.g., series resistor 914) and the switch(e.g., switch 911 a) is reversed, meaning that the plurality of seriesresistors 914 and 915 are electrically connected to the output of thedriver amplifier circuit 902 (common input node) and the inputmultiplexer circuit 910 is inserted between the plurality of seriesresistors 914 and 915 and the plurality of sense circuits 106 and 108.Reversing the order may be advantageous for the design of the switch(e.g., switch 911 a) as the voltage V₂ across the sense circuit may besubstantially lower than the voltage at the output of the driveramplifier circuit 902.

As discussed above, a low frequency disturbance voltage (e.g., at WPTfrequency) may be present at the driver circuit 402 output and at themeasurement amplifier circuit 404 input e.g., due to the voltage inducedinto the sense element (e.g., sense element 107 a) by the WPT magneticfield. If a switch (e.g., switch 911 b) is in open-state, a substantiallow frequency voltage may also be present across the switch. This may beparticularly true during active WPT operation. If too large, theopen-switch voltage may affect any of the switch' open-state electricalcharacteristic or cause damage to the switch. In some implementations,the open-switch voltage is limited by configuring the inductive andcapacitive sense circuits (e.g., sense circuit 106 a and sense circuit108) accordingly, trading-off the open-switch voltage vs. other impacts.

The measurement amplifier circuit 404 is configured to operate as theanalog front-end part of a voltage measurement circuit (e.g., voltagemeasurement circuit 510 as described in connection with FIG. 5A). It isconfigured to selectively (e.g., sequentially) buffer and amplify thevoltage V₂ in each of the plurality of inductive sense circuits 106 andin each of the plurality of capacitive sense circuits 108 and to providea measurement amplifier output voltage signal V_(out) (as indicated inFIG. 9A) based on the respective voltage V₂ at a level suitable forfurther processing e.g., in the signal processing circuit 408 withreference to FIG. 4. In some implementations providing a voltage gain,the voltage V_(out) is larger than V₂. In other implementations, themeasurement amplifier circuit 404 mainly serves for impedance bufferingand is configured for unity gain. In further implementations, the outputvoltage V_(out) is even smaller than V₂. As previously mentioned, a DCflow at any of the plurality of inputs of the measurement amplifiercircuit 404 may be present e.g., caused by a DC offset at theamplifier's 954 input or by certain types of analog switches (e.g.,switch 941 a).

The measurement amplifier circuit 404 includes a transimpedanceamplifier circuit 952, an output multiplexer circuit 940 illustrated inFIG. 9A as a plurality of switches 941 a, 941 b, . . . , 941 n, and aplurality of resistors 944 and 945 (series resistors) connected inseries to the respective output of the output multiplexer circuit 940.The series resistors 944 connected to the inductive sense circuits 106have a resistance R_(ser2) that may differ from the series resistanceR_(ser1), while the series resistors 945 connected to the capacitivesense circuits 108 have a resistance R_(ser4) that may differ fromR_(ser3) and R_(ser2). Further, it includes. The plurality of switches941 a, 941 b, . . . , 941 n are electrically connected to thetransimpedance amplifier circuit's 952 input that is also referred toherein as the common output node of the output multiplexer circuit 940.

The example transimpedance amplifier circuit 952 as illustrated in FIG.9A includes an amplifier 954, a feedback resistor 956 having aresistance R_(f), and a feedback capacitor 958 having a capacitanceC_(f). In some implementations, the amplifier 954 is at least one of alow noise operational amplifier and an amplifier providing highlinearity. The positive input (+) of the amplifier 954 connects toground. Both the feedback resistor 956 and the feedback capacitor 958are electrically connected between the output (V_(out)) and the negativeinput (−) of the amplifier 954. Further, the transimpedance amplifiercircuit 952 is configured to receive an input current I_(in), which isthe output current at the common output node of the output multiplexercircuit 940 and to convert the input current I_(in) into a proportionaloutput voltage V_(out). The conversion gain (transimpedance) isdetermined by the impedance of the parallel connection of the feedbackresistor 956 and the feedback capacitor 958. Since the voltage at thenegative input (−) of the amplifier 954 is virtually zero (virtualground), the transimpedance amplifier circuit 952 presents a virtuallyzero input impedance at its negative input (−).

The transimpedance amplifier circuit 952 together with the seriesresistor (e.g., resistor 944) mimic a voltage measurement circuitcharacteristic at each of the plurality of inputs of the measurementamplifier circuit 404. It may be appreciated that the series resistor(e.g., resistor 944) with a resistance (e.g., R_(ser2)) substantiallylarger (e.g., 10 times larger) than the impedance magnitude |Z₁₁| atseries resonance e.g., of the sense circuit 106 a transforms thevirtually zero impedance input of the transimpedance amplifier circuit952 into a high impedance input satisfying the requirements of a voltagemeasurement circuit 510 as previously specified in connection with FIG.5A. They form together a sign inverting voltage amplifier with an outputvoltage V_(out) proportional to the sign inverse of the voltage V₂.

In an aspect, increasing the resistance (e.g., R_(ser2)) of the seriesresistor (e.g., resistor 944) may improve a voltage measurement circuitcharacteristic but reduce the input current level I. The input currentlevel I_(in) may impact a SNR as previously defined with reference toFIG. 5F. Moreover, increasing the resistance (e.g., R_(ser2)) may reducean impact of the output multiplexer circuit 940 switch (e.g., switch 941a) (e.g., a temperature drift) as discussed above in connection with theinput multiplexer circuit 910. Therefore, in some implementations, theresistances R_(ser2) and R_(ser4) may represent a trade-off between avoltage measurement circuit characteristic, a temperature stability, anda SNR.

In an example implementation variant (not shown herein), an impact ofthe output multiplexer circuit 940 switch (e.g., switch 941 a) (e.g., atemperature drift) is reduced by omitting the output multiplexer circuit940, instead using a plurality (bank) of measurement amplifiers (notshown herein), whose inputs are electrically connected to the respectivemeasurement port (e.g., measurement port 937) of the respective sensecircuit (e.g., sense circuit 106 a) and whose outputs are electricallyconnected to a common output (V_(out)). Each measurement amplifiercircuit 404 is configured to provide a high input impedance and includesan operational amplifier (e.g., amplifier 954) providing a mute controlinput to apply a logic signal indicative for the output MUX controlsignal as indicted in FIG. 9A. The operational amplifier is furtherconfigured to provide a virtually zero gain and a high output impedancewhen muted. An example circuit using a bank of operational amplifiers isdisclosed in U.S. patent application Ser. No. 16/226,156, titled ForeignObject Detection Circuit Using Current Measurement, the entire contentsof which are hereby incorporated by reference.

In an example configuration of the circuit 900, the voltage measurementcircuit characteristic is realized using a resistor (e.g., resistor 944)with a lower resistance (e.g., R_(ser2)) instead using the DC blockcapacitor (e.g., capacitor 929 of sense circuit 106 a) with a higherreactance, together providing a series impedance substantially larger(e.g., 10 times larger) than the impedance of the series circuit e.g.,of the sense element 107 a and the capacitor 504 at series resonance. Inan implementation variant, the series resistor (e.g., resistor 944) isomitted and the voltage measurement circuit characteristic is realizedusing the DC block capacitor (e.g., capacitor 929) configured to providea high enough series impedance. In another implementation variant (notshown herein), the high series impedance as required to mimic a currentsource characteristic is realized at least in part by using at least oneof an inductor (not shown in FIG. 9A) and a leakage inductance of atransformer.

The feedback capacitor 958 provides the transimpedance amplifier circuit952 with a first order low pass filter characteristic to attenuatedisturbance signal components at frequencies higher than the sensefrequency (e.g., high order WPT harmonics). In some implementations, thecapacitance C_(f) may be a trade-off between a reduction in gain at thesense frequency and an attenuation of the high frequency signalcomponents. The feedback capacitor 958 may reduce a risk for signalclipping or non-linear distortion in the amplifier 954 or in the furtherprocessing (e.g., the signal processing circuit 408 with reference toFIG. 4) e.g., during WPT operation. In other words, it may reduce thedynamic range requirements for the measurement amplifier circuit 404 andin the further processing.

In an implementation variant (not shown herein), the transimpedanceamplifier circuit 952 is further enhanced by a supplementary feedbackinductor electrically connected in parallel to the feedback capacitor958 providing the transimpedance amplifier circuit 952 with a bandpasscharacteristic tuned to the sense frequency. This inductor may help tofurther suppress low frequency disturbance signal components (e.g., theWPT fundamental and low frequency harmonics thereof). In such animplementation, the feedback capacitor 958 may be a temperaturecompensation capacitor to compensate for at least a portion of thefeedback inductor's temperature drift.

In another implementation variant (not shown in FIG. 9A), themeasurement amplifier circuit 404 is configured as an active filterproviding a bandpass characteristic similar to the implementationvariant described above using a parallel feedback inductor. This circuitmay not use any inductors. An example implementation of a measurementamplifier circuit 404 using an active filter is illustrated in FIG. 9C.

The output multiplexer circuit 940 including the plurality of switches941 a, 941 b, . . . , 941 n is configured to selectively connect each ofthe plurality of inductive sense circuits 106 and each of the pluralityof capacitive sense circuits 108 via the respective series resistors 944and 945 to the transimpedance amplifier circuit 952 to selectively(e.g., sequentially) buffer and amplify the voltage V₂ at each of theplurality of sense circuits 106 and 108 in response to a current I₀ atthe sense frequency. Therefore, each of the plurality of switches 941 a,941 b, . . . , 941 n is electrically connected to the common output nodethat is electrically connected to the negative input of thetransimpedance amplifier circuit 952. The output multiplexer circuit 940is further configured to receive an output MUX control signal from acontrol circuit (e.g., from the control and evaluation circuit 102 ofFIG. 4) that controls the output multiplexer circuit 940.

As for the input multiplexer circuit 910, each of the plurality ofswitches 941 a, 941 b, . . . , 1041 n may be one of a semiconductoranalog switch (e.g., a single FET switch, a complementary FET switchcomposed of a p-channel and a n-channel type FET), a micro-mechanicalsystems (MEMS) switch or any other type of switch providing asufficiently high current signal attenuation in the open-state. Anexample implementation of an analog switch (e.g., switch 941 a) based ona single FET is illustrated in FIG. 9D.

As previously discussed in connection with the input multiplexer circuit910, each of the plurality of switches 941 a, 941 b, . . . , 941 n mayexhibit a closed-state series resistance, an equivalent open-stateseries capacitance, and an equivalent parallel capacitance on each sideof the switch. It may be appreciated that the closed-state seriesresistance (e.g., of switch 941 a) may be non-critical for thefunctioning of the measurement amplifier circuit 404 since it mergesinto an overall series resistance R_(ser2) or R_(ser4). It may also beappreciated that the total capacitive load produced by the plurality ofparallel capacitances at the common output node of the outputmultiplexer circuit 940 may be non-critical since it is in parallel tothe virtually zero-impedance input of the transimpedance amplifiercircuit 952.

An example output multiplexer circuit 940 for a sense frequency of 3 MHzmay use complementary FET switches with the same characteristics asthose used for the example input multiplexer circuit 910 as specifiedabove.

As previously discussed with reference to the input multiplexer circuit910, a low frequency voltage (e.g., at WPT frequency) may also bepresent across the switch (e.g., switch 941 a) when WPT is active andthe switch in open-state. If too large, this open-switch voltage mayaffect any of the switch's open-state electrical characteristics orcause damage to the switch. In some implementations, the open-switchvoltage is limited as previously discussed for the input multiplexercircuit 910.

In an implementation variant (not shown herein), the order of the seriesresistor (e.g., resistor 944) and the switch (e.g., switch 941 a) isreversed, meaning that the series resistor (e.g., resistor 944) iselectrically connected to the input of the transimpedance amplifiercircuit 952 (common output node) and the output multiplexer circuit 940is inserted between the plurality of sense circuits 106 and 108 and theplurality of resistors 944 and 945.

A further implementation variant of the circuit 900 (not shown herein)omits the output multiplexer circuit 940 e.g., for reasons of complexityand cost reduction. In theory, the input multiplexer circuit 910 may beenough to selectively (e.g., sequentially) drive a sense circuit (e.g.,sense circuit 106 a) with the current I₁ and to selectively measure thevoltage V₂ at its measurement port 937 in response to the current I₁.Because the measurement amplifier circuit 404 is configured as a voltagesummation amplifier, its output voltage V_(out) is indicative of the sumof the voltages at each of the plurality of measurement ports 937. Sincethe voltages V₂ at the inactive sense circuits (not driven by thecurrent are ideally zero, the output voltage V_(out) is indicative ofthe voltage V₂ of the active sense circuit (e.g., sense circuit 106 a)driven by the current I₁. However, in practice, disturbance voltagese.g., inductively or capacitively coupled into the sense element (e.g.,inductive sense element 107 b) may also sum up resulting in a lower SNRas compared to a circuit 900 using the output multiplexer circuit 940.Moreover, a circuit 900 omitting the output multiplexer circuit 940 maynot support supplementary inter-sense circuit transimpedance Z₁₂measurements as described above. The inter-sense circuit transimpedanceZ₁₂ between the measurement port 936 of sense circuit 106 a and themeasurement port 937 of sense circuit 106 b may be measured by settingthe switches 911 a and 941 b to the close-state and the other switchesof the plurality of switches 911 a, 911 b, . . . , 911 n to theopen-state.

A numerical specification and some resulting performance figures of anexample circuit 900 with respect to the inductive sense circuits 106configured for a nominal sense frequency ƒ_(s)=3 MHz are given in TABLE7. A WPT operating frequency ƒ_(WPT)=85 kHz is assumed. TABLE 7 alsoincludes a system Q-factor defined as the ratio

Q _(s;ys) ≈|ΔV _(out) ′|/|ΔZ _(r)′|  (352)

where |ΔV_(out)′| denotes the magnitude fractional change of themeasurement amplifier circuit's 404 output voltage V_(out) caused by anobject (e.g., object 110) and |ΔZ_(r)′| the magnitude normalizedreflected impedance of the object as defined by Equations (6) an (211)for an inductive sense element (e.g., inductive sense element 107 a) anda capacitive sense element (e.g., capacitive sense element 109 n),respectively. Further, TABLE 7 includes a quality of the measurementcircuit 104 defined as the ratio:

Q _(mc) ≈|ΔV _(out) ′|/|ΔI ₁′|  (353)

where |ΔV_(out)′| denotes the magnitude fractional change of themeasurement amplifier circuit's 404 output voltage V_(out) caused by anobject (e.g., object 110) and |ΔI₁′| the magnitude fractional change ofthe driver circuit's 402 output current I₁ caused by that object.Moreover, TABLE 7 includes a degradation of the fractional change in themeasurement amplifier circuit's 404 output voltage V_(out) defined as:

Y _(Vout)≈1−(|ΔV _(out) ′|/|ΔZ′|)  (354)

where |ΔZ′| denotes the magnitude fractional change of the intra-sensecircuit transimpedance Z_(2a1a) as defined by Equation (350) for thesense circuit 106 a by example. This degradation may be the result ofimperfections in the driver circuit 402 and the measurement amplifiercircuit 404 and in other circuit elements as needed in a practicalimplementation.

TABLE 7 Item Symbol Value Remarks Input multiplexer circuit 910 switchclosed- 5 Ω state resistance Series resistor 914 resistance R_(ser1) 100Ω DC block capacitor 928 capacitance C_(b1) 3.3 nF Inductive senseelement 107a inductance L 5 pH Inductive sense element 107a equivalent R3.8 Ω Q-factor Q = 25 series resistance Capacitor 504 capacitance C_(s)563 pF Parallel inductor 506 inductance L_(p) 18 μH DC block capacitor929 capacitance C_(b2) 3.3 nF Series resistor 944 resistance R_(ser2)100 Ω Output multiplexer circuit 940 switch 5 Ω closed-state resistanceFeedback resistor 956 resistance R_(f) 560 Ω Feedback capacitor 958capacitance C_(f) 100 pF Sense circuit 106a parallel resonance 1.4 MHzExact sense frequency 3.0006 MHz Transimpedance (minimum magn.) Z₂₁₀3.769 Ω Meas. port 936 to 937 System Q-factor Q_(sys) 23.4 Quality ofmeasurement circuit 104 Q_(mc) 29.2 Residual angle error ε −0.6°Degradation of fractional change at meas. γV_(out) 6.5% circuit 104output (V_(out)) Driver amplifier circuit 902 output voltage 7.9 V_(pp)2.8 V_(rms) Driver amplifier circuit 902 output current 25.5 mA_(rms)Driver amplifier circuit 902 output power 71.4 mW Driver circuit 402output current 25.5 mA_(rms) Inductive sense element 107a current I_(L)24.6 mA_(rms) (sense current) Transimpedance amplifier circuit 952 inputI_(in) 0.87 mA_(rms) 3 MHz sense signal current Measurement amplifiercircuit 404 output V_(out) 0.95 V_(pp) 3 MHz sense signal, voltage 0.34V_(rms) WPT fundamental disturbance voltage V_(sW) 30 V_(rms) 85 kHzcomponent induced into inductive sense element 107a WPT fundamentaldisturbance voltage at 0.25 V_(pp) 85 kHz component measurementamplifier circuit 404 output SNR with respect to WPT fundamental SNR_(W)11.7 dB 85 kHz component disturbance voltage WPT fundamental disturbancevoltage 8.2 V_(pp) 85 kHz component across input multiplexer circuit 910switch when open WPT fundamental disturbance voltage 0.2 V_(pp) 85 kHzcomponent across output multiplexer circuit 940 switch when open

A numerical specification of an example circuit 900 with respect to thecapacitive sense circuits 108 configured for a nominal sense frequencyƒ_(s)=3 MHz are given in TABLE 8. A WPT operating frequency ƒ_(WPT)=85kHz is assumed. TABLE 8 also includes the system Q-factor Q_(sys) asdefined above by Equation (352), the quality of the measurement circuit104 as defined by Equation (353), and the degradation of the fractionalchange in the measurement amplifier circuit's 404 output voltage V_(out)as defined by Equation (354).

TABLE 8 Item Symbol Value Remarks Input multiplexer circuit 910 switchclosed- 5 Ω state resistance Series resistor 915 resistance R_(ser3) 100Ω DC block capacitor 930 capacitance C_(b3) 3.3 nF Inductive senseelement 109n capacitance C 30 pF Series inductor 724 inductance L_(s) 94uH includes transformer 726 leakage inductance Series inductor 724equivalent series R_(Ls) 118 Ω Q-factor Q_(Ls) = 15, resistance includestransf. 726 losses Transformer 726 secondary-referred main L_(m) 270 uHinductance Transformer 726 voltage transformation 1:n_(VT) 1:5 ratio DCblock capacitor 931 capacitance C_(b4) 3.3 nH Series resistor 945resistance R_(ser4) 100 Ω Output multiplexer circuit 940 switch closed-5 Ω state resistance Feedback resistor 956 resistance R_(f) 470 ΩFeedback capacitor 958 capacitance C_(f) 120 pF Sense circuit 108aparallel resonance 1.52 MHz Exact sense frequency 3.0023 MHzTransimpedance (minimum magn.) Z₂₁₀ 4.713 Ω Meas. port 938 to 939 SystemQ-factor Q_(sys) 13.8 Quality of measurement circuit 104 Q_(mc) 23.5Residual angle error ε  −0.9° Degradation of fractional change at meas.8% circuit 104 output (V_(out)) Driver amplifier circuit 902 outputvoltage 7.92 V_(pp) 2.8 V_(rms) Driver amplifier circuit 902 outputcurrent 25.3 mA_(rms) Driver amplifier circuit 902 output power 70.8 mWDrive circuit 402 output current 25.3 mA_(rms) Capacitive sense element109a current I_(c) 4.8 mA_(rms) (sense current) Transimpedance amplifiercircuit 952 input I_(in) 1.1 mA_(rms) 3 MHz sense signal currentMeasurement amplifier circuit 404 output V_(out) 0.98 V_(pp) 3 MHz sensesignal voltage 0.35 V_(rms) WPT fundamental disturbance voltage V_(sW)150 V_(rms) 85 kHz component induced into capacitive sense element 109aWPT fundamental disturbance voltage at 0.16 V_(pp) 85 kHz componentmeasurement amplifier circuit 404 output SNR with respect to WPTfundamental SNR_(W) 15.6 dB 85 kHz component disturbance voltage Voltageacross input multiplexer circuit 910 8.1 V_(pp) 85 kHz component switchwhen open and WPT active Voltage across output multiplexer circuit 9400.2 V_(pp) 85 kHz component switch when open and WPT active

FIG. 9C is a circuit diagram illustrating an example implementation of ameasurement amplifier circuit 404 using an active filter providing abandpass characteristic as previously mentioned with reference to FIG.9A. The measurement amplifier circuit 404 is configured to selectively(e.g., sequentially) buffer and amplify the voltage V₂ in each of theplurality of inductive sense circuits 106 and in each of the pluralityof capacitive sense circuits 108 and to provide an output voltageV_(out) based on the respective voltage V₂ at a level suitable forfurther processing e.g., in the signal processing circuit 408 withreference to FIG. 4. In some implementations, the bandpass filtercharacteristic of the active filter substantially equals the bandpasspass characteristics of the measurement amplifier circuit 404 based on atransimpedance amplifier circuit 952 including a feedback networkcomprising a resistor, a capacitor, and an inductor as previouslydescribed in connection with FIG. 9A. However, the temperature stabilityof the measurement amplifier circuit 404 of FIG. 9C using an activefilter may be higher than that of a measurement amplifier circuit 404using a feedback inductor.

The example measurement amplifier circuit 404 of FIG. 9C includes atransimpedance amplifier circuit 952, an output multiplexer circuit 940including a plurality of switches 941 a, 941 b, . . . , 941 n, and aplurality of capacitors (series capacitors 946 and 947) connected inseries to the respective switch. In the measurement amplifier circuit404 of FIG. 9C, the resistors 944 and 945 with reference to FIG. 9A aresupplanted by respective series capacitors 946 and 947 havingcapacitance C₃ and C₄ configured to provide the high input impedance andthe active filter characteristic as required in some implementations ofthe measurement amplifier circuit 404. The transimpedance amplifiercircuit includes an amplifier 954 (e.g., at least one of a low noiseoperational amplifier and an operational amplifier providing highlinearity). The amplifier's 954 positive input (+) is electricallyconnected to ground. A feedback capacitor 958 with capacitance C₁ iselectrically connected between the output (V_(out)) and the negativeinput (−) of the amplifier 954. Further, a feedback resistor 956 withresistance R₁ is electrically connected between the output (V_(out)) andthe input (I_(in)) of the transimpedance amplifier circuit 952. Aresistor 959 with resistance R₂ is electrically connected between theinput (I_(in)) of the transimpedance amplifier circuit 952 and thenegative input (−) of the amplifier 954. A capacitor 960 withcapacitance C₂ is electrically connected between the input (I_(in)) ofthe transimpedance amplifier circuit 952 and ground. Further, each ofthe capacitors 946 having a capacitance C₃ capacitively couples arespective switch (e.g., switch 941 a) to the respective inductive sensecircuit (e.g., inductive sense circuit 106 a), while each of thecapacitors 947 having a capacitance C₄ capacitively couples a respectiveswitch (e.g., switch 941 n) to the respective capacitive sense circuit(e.g., capacitive sense circuit 108 a). The second terminal of theplurality of switches 941 a, 941 b, . . . , 941 n is connected to thecommon output node corresponding to the input (I_(in)) of thetransimpedance amplifier circuit 952.

Active filters providing a bandpass characteristic may be implemented ina number of ways which may be well known to those skilled in the art.However, not many of these implementations may satisfy the requirementsof a voltage measurement circuit (e.g., voltage measurement circuit510).

FIG. 9D is a circuit diagram illustrating an example implementation ofan analog switch (e.g., 941 a) as used in an example implementation ofthe output multiplexer circuit 940. The analog switch uses a singlefield-effect transistor (FET) 942 (e.g., a n-channelmetal-oxide-semiconductor field-effect transistor (MOSFET)). The analogswitch is controlled by a switch control signal applied to the gate (G)of the FET 942. The Drain (D) of the FET 942 is DC biased with apositive voltage +V_(B) via a resistor having a resistance R_(B1).(e.g., in the kΩ range). The positive DC voltage +V_(B) ensures that thevoltage across the FET channel remains positive at any time when inopen-state and in presence of a superimposed alternating voltage acrossthe FET channel (e.g., during WPT operation as previously discussed withreference to FIG. 9). Further, a negative DC voltage −V_(B) is appliedat the Source (S) of the FET 942. This voltage may be adjusted tosubstantially compensate for any DC at the input of the transimpedanceamplifier circuit 952 caused by the positive DC voltage +V_(B) and toreduce a DC offset at the output (V_(out)).

FIG. 10 is a circuit diagram of a circuit 1000 illustrating anotherexample implementation of a portion of a multi-purpose detection circuit100. The circuit 1000 of FIG. 10 illustrates an analog front-end circuitportion of the multi-purpose detection circuit 100 with reference toFIGS. 1 and 4. FIG. 10 excludes various other signal generation,processing, control and evaluation circuits (e.g., as shown in FIG. 4)that may be needed in some implementations of a multi-purpose detectioncircuit 100. The circuit 1000 implements inductive and capacitivesensing by measuring an impedance based on the current source voltagemeasurement approach as previously described in connection with FIG. 5C.

As with the circuit 900 of FIG. 9A, the circuit 1000 may be subdividedinto a driver circuit 402, a plurality of inductive sense circuits 106including sense circuits 106 a, 106 b, . . . , 106 n (106 n not shown inFIG. 10 for purposes of illustration), a plurality of capacitive sensecircuits 108 including sense circuits 108 a, 108 b, . . . , 108 n (108 aand 108 b not shown in FIG. 10 for purposes of illustration), and ameasurement amplifier circuit 404 with reference to the generic blockdiagram of FIG. 4. However, each of the plurality of inductive sensecircuits 106 a, 106 b, . . . , 106 n is based on the sense circuit 541of FIG. 5C, while each of the plurality of capacitive sense circuits 108a, 108 b, . . . , 108 n is based on the sense circuit 721 of FIG. 7C.The dots indicated in FIG. 10 shall indicate that the number ofinductive sense circuits 106 and/or the number of capacitive sensecircuits 108 may be greater than three as previously noted withreference to FIG. 1.

In the example implementation shown in FIG. 10, each of the plurality ofinductive sense circuits 106 have an identical circuit topology.Likewise, each of the plurality of capacitive sense circuits 108 have anidentical circuit topology. Therefore, descriptions given below for theinductive sense circuit 106 a also apply to the other inductive sensecircuits (e.g., 106 b) and descriptions given below for the capacitivesense circuit 108 n also apply to the other capacitive sense circuits(e.g., 108 a).

Each of the plurality of inductive sense circuits 106 provides a firstmeasurement port 936 (indicated in FIG. 10 by a terminal) for applyingan electrical voltage V₁ (as indicated in FIG. 10) and a secondmeasurement port 937 (indicated in FIG. 10 by a terminal) for measuringan electrical current I₂ (as indicated in FIG. 10) e.g., in response tothe voltage V₁. Likewise, each of the plurality of capacitive sensecircuits 108 provides a first measurement port 938 (indicated in FIG. 10by a terminal) for applying the voltage V₁ (as indicated in FIG. 10) anda second measurement port 939 (indicated in FIG. 10 by a terminal) formeasuring the current I₂ (as indicated in FIG. 10) e.g., in response tothe current V₁. Though providing two ports, the sense circuits 106 and108 as illustrated in FIG. 10 may be considered as one-port circuits asfurther discussed below.

The driver circuit 402 includes an input multiplexer circuit 910 toselectively (e.g., sequentially) apply the voltage V₁ to each of theplurality of sense circuits 106 and 108. Likewise, the measurementamplifier circuit 404 includes an output multiplexer circuit 940configured to selectively (e.g., sequentially) measure the current I₂ ineach of the plurality of sense circuits 106 and 108. More specifically,but not indicated in FIG. 10 for purposes of illustration, the voltageV₁ applied to the sense circuit 106 a may be denoted by V_(1a), thevoltage V₁ applied to the sense circuit 106 b may be denoted as V_(1b),etc. Likewise, the currents I₂ in the sense circuits 106 a and 106 b maybe denoted by I_(2a) and I₂ b, respectively.

The circuit 1000 may be configured and operated in a mode to selectively(e.g., sequentially) measure the intra-sense circuit transimpedance Z₂₁e.g., between the measurement ports 936 and 937 of each of the pluralityof the sense circuits 106 defined as:

Z _(2a1a) ≈V _(1a) /I _(2a)  (355)

As mentioned above and further discussed below, the two-porttransimpedance Z₂₁ substantially equals the one-port impedance Z₁₁ as itmay be measured at the first measurement port (e.g., measurement port936) with the second measurement port (e.g., measurement port 937)short-circuited.

However, as opposed to the circuit 900 of FIG. 9A, the circuit 1000 asillustrated in FIG. 10 may not support measuring the inter-sense circuittransimpedance Z₁₂ as described with reference to FIG. 9A. Measuring theinter-sense circuit transimpedance Z₁₂ may require a different inputmultiplexer circuit 910 as discussed in greater detail below.

The inductive sense circuit 106 a includes an inductive sense element107 a including a sense coil (e.g., sense coil 502 of FIG. 5A) with aninductance L, a first capacitor (e.g., capacitor 544 of FIG. 5C) withcapacitance C_(p), a second capacitor (e.g., capacitor 546 of FIG. 5C)with a capacitance C_(s). As opposed to the sense circuits 106 of thecircuit 900 of FIG. 9A, the sense circuits 106 of the circuit 1000 maynot require an inductor (e.g., inductor 506) other than the inductivesense element (e.g., inductive sense element 107 a) since the capacitor546 may suffice to provide the required high pass filter characteristic.The first capacitor (parallel capacitor 544) is electrically connectedin parallel to the inductive sense element 107 a. The parallel circuitof capacitor 544 and inductive sense element 107 a is capacitivelycoupled to the measurement port 936 via capacitor 546 and alsoelectrically connected to the measurement port 936. None of thecomponents of the sense circuit 106 a is ground-connected, meaning thatthere is ideally no current flow towards ground when the voltage V₁ isapplied at the measurement port 936. Therefore, the sense circuit 106 amay be considered as a one-port rather than a two-port circuit. Inpractice however, some residual ground current flow may exist due toparasitic capacitances of the inductive sense element 107 a e.g., asillustrated in FIG. 5A.

In an example implementation of the circuit 1000 of FIG. 10, each of theplurality of inductive sense circuits 106 is configured to provide aminimum of the impedance magnitude |Z₁₁| (series resonance)substantially at the sense frequency as previously discussed withreference to the circuit 540 of FIG. 5C.

In another aspect, the capacitors 544 and 546 may be configured toprovide an impedance magnitude |Z₁₁| at the sense frequency in asuitable range for the measurement circuit (e.g., measurement circuit104 of FIG. 4) by adjusting the capacitance ratio n_(C)=C_(p)/C_(s)accordingly as previously discussed with reference to FIG. 5G. Byincreasing the ratio n_(C), the impedance |Z₁₁| may be increased to alevel substantially higher than that of the inductive sense circuit 106a of the circuit 900 of FIG. 9A. A higher impedance |Z₁₁| may be arequirement of the voltage source current measurement technique e.g.,for selectively applying the voltage V₁ to each of the plurality ofsense circuits 106 by the driver circuit 402 without exceeding an outputcurrent limit. It may also be a requirement for selectively measuring acurrent in each of the plurality of sense circuits 106 by themeasurement amplifier circuit 404 without exceeding an input currentlimit. Further, it may be a requirement to reduce an impact of the inputmultiplexer circuit 910 and the output multiplexer circuit 940 asfurther discussed below.

In some implementations, the capacitors 544 and 546 are of a type with alow temperature coefficient providing high thermal stability (e.g., aNP0-type capacitor) reducing thermal drift of an electricalcharacteristic (e.g., an impedance) as measured at each of the pluralityof inductive sense circuits 106 a, 106 b, . . . , 106 n.

Moreover, as previously discussed in connection with FIG. 5C, theinductive sense circuit 106 a in conjunction with the voltage sourcecurrent measurement technique creates a high pass filter characteristicto attenuate a low frequency disturbance component in the current I₂ forpurposes as previously discussed with reference to FIG. 9A. The seriescapacitor 546 together with the parallel capacitor 544 may be configuredto attenuate this low frequency disturbance component to a level e.g.,below the level of the current I₂ in response to the respective sensevoltage V₁ at the sense frequency, while maintaining series resonance atthe sense frequency as previously discussed with reference to FIG. 5G.At the sense frequency, this high pass filter characteristic may exert aminor impact on the current I₂ and thus on the measurement of theimpedance Z₁₁ and which may be corrected in a further processing (e.g.,in the signal processing circuit 408 of FIG. 4) as previously discussedwith reference to the circuit 900 of FIG. 9A.

In another aspect, the inductive sense circuits 106 may not need anysupplementary capacitors (e.g., capacitors 928 and 929) for purposes ofDC blocking as previously discussed with reference to the circuit 900 ofFIG. 9A, as the series capacitor 546 already blocks any DC. Therefore,the passive component count of the plurality of inductive sense circuit106 a of FIG. 10 may be lower as compared to the circuit 900 of FIG. 9A.

The capacitive sense circuit 108 n includes a capacitive sense element109 n including a sense electrode (e.g., sense electrode 702 of FIG. 7Cillustrating a single-ended sense electrode) with a capacitance C, aseries inductor 724 (e.g., series inductor 724 of FIG. 7C) with aninductance L_(s), a transformer 726 (e.g., transformer 726 of FIG. 7C)providing a primary and secondary port, a secondary-referred maininductance L_(m) and a voltage transformation ratio 1:n_(VT). Further,it includes a capacitor 930 with a capacitance C_(b) e.g., for purposesas previously discussed with reference to the sense circuits 106 and 108of FIG. 9A. The inductor 724 is electrically connected in series to thecapacitive sense element 109 n. The series circuit of inductor 724 andcapacitive sense element 109 n is electrically connected to thetransformer's 726 secondary port that also connects to ground. Further,the transformer's 726 primary port is capacitively coupled to themeasurement port 938 via capacitor 930 and also electrically connects tothe measurement port 939. Though the transformer's 726 secondary portelectrically connects to ground, the sense circuit 108 n ideally may notbe ground-related. In an implementation using a transformer 726 composedof a primary winding and a galvanically isolated secondary winding aspreviously described with reference to FIG. 9A, there is ideally nocurrent flow towards ground when the voltage V₁ is applied at themeasurement port 938. Therefore, the capacitive sense circuits 108 n maybe considered a one-port rather than a two-port circuit. In practicehowever, some residual ground current flow may exist due to a parasiticinterwinding capacitance of the transformer 726.

In an implementation variant using a capacitive sense element 109 nincluding a double-ended sense electrode (not shown herein), the sensecircuits 108 may not need a transformer (e.g., transformer 726) forpurposes of ground-decoupling. In another implementation variant using adouble-ended sense electrode, each of the plurality of capacitive sensecircuits 108 is based on the sense circuit 781 as illustrated in FIG.7I.

In an example implementation of the circuit 1000 of FIG. 10, each of theplurality of capacitive sense circuits 108 is configured to provide aminimum of the impedance magnitude |Z₁₁| (series resonance)substantially at the sense frequency as previously discussed withreference to the circuit 720 of FIG. 7C.

In another aspect, the transformer 726 may be configured to provide animpedance |Z₁₁| at the sense frequency in a suitable range for themeasurement circuit (e.g., measurement circuit 104 of FIG. 4) byadjusting the voltage transformation ratio 1:n_(VT) accordingly, where nrefers to the transformer's 726 secondary-side. By decreasing n, theimpedance |Z₁₁| may be increased to a level higher than that of thesense circuit 108 n of the circuit 900 of FIG. 9A. A higher impedance|Z₁₁| may be a requirement of the voltage source current measurementtechnique as previously discussed.

The secondary-referred main inductance L_(m) of the transformer 726together with the capacitance C of the capacitive sense element 109 nform a 2^(nd) order high pass filter to attenuate a low frequencydisturbance component in the current I₂ as previously discussed withreference to FIG. 7C. The transformer 726 may be configured to attenuatethis low frequency disturbance component to a level e.g., significantlybelow the level of the current I₂ in response to the respective voltageV₁ at the sense frequency. At the sense frequency, this high pass filtermay exert a minor impact on the current I₂ and thus on the measurementof the impedance Z₁₁ and which may be corrected in a further processing(e.g., in the signal processing circuit 408 of FIG. 4) as previouslydiscussed with reference to FIG. 7A.

The driver circuit 402 includes a driver amplifier circuit 902, an inputmultiplexer circuit 910 illustrated in FIG. 10 as a plurality ofswitches 911 a, 911 b, . . . , 911 n, and a plurality of resistors 1014and 1015 (parallel resistors) connected in parallel to the respectiveoutput of the input multiplexer circuit 910 for purposes as describedbelow. The parallel resistors 1014 connected to the inductive sensecircuits 106 have a resistance R_(par1); while the parallel resistors1015 connected to the capacitive sense circuits 108 have a resistanceR_(par3) that may generally differ from R_(par1).

Further, the driver circuit 402 is configured to operate as a voltagesource (e.g., voltage source 552 as described in connection with FIG.5C) and to selectively (e.g., sequentially) apply a voltage signal V₁ atthe sense frequency to each of the plurality of inductive sense circuits106 and to each of the plurality of the capacitive sense circuits 108.The voltage signal V₁. (e.g., a sinusoidal sense signal) is based on adriver input signal which may be an output of the signal generatorcircuit 406 with reference to FIG. 4. The driver amplifier circuit 902as illustrated in FIG. 10 by example includes an amplifier 904 andexternal resistance circuitry comprising a first (feedback) resistor 906and a second resistor 908 for adjusting a gain. In some implementations,the amplifier 904 is at least one of a low noise operational amplifierand an operational amplifier providing high linearity. The driveramplifier circuit 902 is configured to receive the driver input signaland to provide a corresponding output with an accurate and stablevoltage (a voltage source output).

The input multiplexer circuit 910 includes a plurality of switches 911a, 911 b, . . . , 911 n and is configured to selectively connect each ofthe plurality of inductive sense circuits 106 and each of the pluralityof capacitive sense circuits 108 to the driver circuit 402 toselectively (e.g., sequentially) apply the voltage V₁ at the sensefrequency to each of the plurality of inductive sense circuits 106 andto each of the plurality of capacitive sense circuits 108. Therefore,each of the plurality of switches 911 a, 911 b, . . . , 911 n iselectrically connected to the driver amplifier circuit's 902 output thatis also referred to as the common input node. The input multiplexercircuit 910 is further configured to receive an input MUX control signalfrom a control circuit (e.g., from the control and evaluation circuit102 of FIG. 4) that controls the input multiplexer circuit 910.

Each of the plurality of switches 911 a, 911 b, . . . , 911 n may be oneof a type as previously mentioned with reference to the circuit 900 ofFIG. 9A. It may be appreciated that the closed-state resistance of theswitch (e.g., switch 911 a) may be less critical if the impedancemagnitude |Z₁₁| at series resonance e.g., of the sense circuit 106 a issubstantially higher (e.g., if the capacitance ratio n_(C) issufficiently large as previously discussed). A high enough impedancemagnitude |Z₁₁| may also reduce an impact of the switch' (e.g., switch911 a) temperature dependent closed-state resistance and thus improve atemperature stability of the driver circuit 402 as previously discussedwith reference to the circuit 900 of FIG. 9A. It may also be appreciatedthat the total capacitive load produced by the plurality of parallelcapacitances at the common input node may be non-critical since it is inparallel to the voltage source output of the driver amplifier circuit902.

The switch (e.g., switch 911 a) of an example input multiplexer circuit910 may use complementary FET switches with a closed-state resistance of5 Ω, an equivalent open-state series capacitance of 3 pF (correspondingto a series reactance of 17.71 k∩ at a sense frequency of 3 MHz), and anequivalent parallel capacitance of 12 pF on each side of the switch.

While in the circuit 900 series resistors 914 and 915 are used totransform the amplifier's 904 voltage source output into a currentsource output, parallel resistors 1014 and 1015) are employed in thecircuit 1000 to limit the open-switch voltage (e.g., at WPT frequency)across the input multiplexer circuit 910 switch (e.g., switch 911 a). Itmay be appreciated that decreasing the resistance (e.g., R_(par1)) ofthe parallel resistor (e.g., parallel resistor 1014) may reduce theopen-switch voltage. However, it will also increase an output current ofthe driver amplifier circuit 902 as more current will be diverted toground. This may increase a voltage drop across the switch (e.g., switch911 a) and hence increasing an impact of the switch' as previouslydiscussed with reference to FIG. 9A. Therefore, in some implementations,the resistances R_(par1) and R_(par3) are a trade-off between a driveramplifier circuit 902 output load, temperature stability, and theopen-switch voltage. In an example implementation, the parallel resistor(e.g., parallel 1014) provides a resistance (e.g., R_(par1)) in theorder of the impedance |Z₁₁|.

Therefore, in an implementation variant (not shown herein), the inputmultiplexer circuit 910 is incorporated into the driver amplifiercircuit 902 also employing an additional (third) multiplexer circuit ina feedback path. This implementation variant may provide a regulated(stable) voltage source characteristic at each of the plurality ofoutputs of the driver circuit 402 substantially eliminating the effectof the switch' temperature dependent closed-state resistance. An examplecircuit of the driver circuit 402, which is voltage regulated, isdisclosed in U.S. patent application Ser. No. 16/226,156, titled ForeignObject Detection Circuit Using Current Measurement.

The measurement amplifier circuit 404 is configured to operate as theanalog front-end part of a current measurement circuit (e.g., currentmeasurement circuit 550 as described in connection with FIG. 5C). It isconfigured to selectively (e.g., sequentially) buffer and convert thecurrent I₂ in each of the plurality of inductive sense circuits 106 andin each of the plurality of capacitive sense circuits 108 and to providean output voltage signal V_(out) (as indicated in FIG. 9A) based on therespective current I₂ at a level suitable for further processing e.g.,in the signal processing circuit 408 with reference to FIG. 4. Themeasurement amplifier circuit 404 includes a transimpedance amplifiercircuit 952, an output multiplexer circuit 940 illustrated in FIG. 10 asa plurality of switches 941 a, 941 b, . . . , 941 n, and a plurality ofresistors 1044 and 1055 (parallel resistors) connected in parallel tothe respective input of the output multiplexer circuit 940. The parallelresistors 1044 connected to the inductive sense circuits 106 have aresistance R_(par2) that the capacitive sense circuits 108 have aresistance R_(par4) that generally differs from R_(par3) and R_(par2).The plurality of switches 941 a, 941 b, . . . , 941 n are electricallyconnected to the transimpedance amplifier circuit's 952 input that isalso referred to herein as the common output node.

As with the parallel resistor 1014, decreasing the resistance (e.g.,R_(par2)) of the parallel resistor 1044 will reduce the open-switchvoltage (e.g., at WPT frequency) the output multiplexer circuit 940switch (e.g., switch 941 a). As opposed to the parallel resistor 1014,decreasing the resistance of the parallel resistor 1044 may be lesscritical since the voltage across the parallel resistor 1044 is low whenthe switch (e.g., switch 941 a) is in closed-state. In an exampleimplementation, the resistances R_(par2) and R_(par4) substantiallymatch the respective impedance |Z₁₁| at series resonance and aresubstantially higher than the switch' closed-state resistance.

The example transimpedance amplifier circuit 952 as illustrated in FIG.10 includes an amplifier 954, a feedback resistor 956 having aresistance Rf, and a feedback capacitor 958 having a capacitance C_(f).In some implementations, the amplifier 954 is at least one of a lownoise operational amplifier and an amplifier providing high linearity.The positive input (+) of the amplifier 954 connects to ground. Both thefeedback resistor 956 and the feedback capacitor 958 are electricallyconnected between the output (V_(out)) and the negative input (−) of theamplifier 954. Further, the transimpedance amplifier circuit 952 isconfigured to receive an input current I_(in), which is the outputcurrent at the common output node of the output multiplexer circuit 940and to convert the input current I_(in) into a proportional outputvoltage V_(out). The conversion gain (transimpedance) is determined bythe impedance of the parallel connection of the feedback resistor 956and the feedback capacitor 958. Since the voltage at the negative input(−) of the amplifier 954 is virtually zero (virtual ground), thetransimpedance amplifier circuit 952 presents a virtually zero inputimpedance at its negative input (−). The feedback capacitor 958 providesthe transimpedance amplifier circuit 952 with a first order low passfilter characteristic to attenuate disturbance signal components atfrequencies higher than the sense frequency (e.g., high order WPTharmonics) as previously discussed with reference to FIG. 9A.

In other implementation variants (not shown herein), the filtering ofthe transimpedance amplifier circuit 952 is further enhanced in similarways as previously described with reference to FIG. 9A.

In a further implementation variant (not shown herein), thetransimpedance amplifier circuit 952 additionally includes an inputtransformer e.g., for purposes of transforming an input current I_(in).The transformer may be used e.g., to reduce the current to a level notexceeding an input current constraint of the amplifier 954 and henceallowing the drive current I₁ and eventually the sense currents I_(L)and I_(C) in the respective sense elements 107 a, 107 b, 107 n and 109a, 109 b, . . . , 109 n to be increased. An example transimpedanceamplifier circuit 952 using an input transformer is disclosed in U.S.patent application Ser. No. 16/226,156, titled Foreign Object DetectionCircuit Using Current Measurement.

The output multiplexer circuit 940 including the plurality of switches941 a, 941 b, 941 n is configured to selectively connect each of theplurality of inductive sense circuits 106 and each of the plurality ofcapacitive sense circuits 108 to the transimpedance amplifier circuit952 to selectively (e.g., sequentially) buffer and convert the currentI₂ at each of the plurality of sense circuits 106 and 108 in response tothe voltage V₁ at the sense frequency. Therefore, each of the pluralityof switches 941 a, 941 b, . . . , 941 n is electrically connected to thecommon output node that is electrically connected to the negative inputof the transimpedance amplifier circuit 952. The output multiplexercircuit 940 is further configured to receive an output MUX controlsignal from a control circuit (e.g., from the control and evaluationcircuit 102 of FIG. 4) that controls the output multiplexer circuit 940.

As with the input multiplexer circuit 910, each of the plurality ofswitches 941 a, 941 b, . . . , 1041 n may be one of a type of switch aspreviously specified with reference to the output multiplexer circuit940 of the circuit 900 of FIG. 9A. It may be appreciated that theclosed-state series resistance (e.g., of switch 941 a) may benon-critical for the functioning of the measurement amplifier circuit404 if the impedance |Z₁₁| of both the sense circuits 106 and 108 issubstantially higher than the closed-state series resistance of theswitch (e.g., switch 911 a). It may also be appreciated that the totalcapacitive load produced by the plurality of parallel capacitances atthe common output node of the output multiplexer circuit 940 may benon-critical since it is in parallel to the virtually zero-impedanceinput of the transimpedance amplifier circuit 952.

An example output multiplexer circuit 940 for a sense frequency of 3 MHzmay use complementary FET switches with the same characteristics asthose used for the example input multiplexer circuit 910 as specifiedabove.

As mentioned above, the circuit 1000 as illustrated in FIG. 10 may notsupport measuring the inter-sense circuit transimpedance Z₁₂ asdescribed with reference to FIG. 9A. However, in an implementationvariant (not shown herein), the circuit 1000 is equipped with an inputmultiplexer circuit 910 including a plurality of tri-state switches. Anexample tri-state switch (not shown herein) may be controlled to one ofa first state that is an open state, a second state that is a closedstate to connect a sense circuit (e.g., sense circuit 106 a) e.g., tothe output of the driver amplifier circuit 902, and a third state thatis also a closed-state to shorten the sense circuit at the measurementport (e.g., measurement port 936) to ground. For example, thetransimpedance Z₁₂ between the measurement port 936 of sense circuits106 a and measurement port 937 of sense circuit 106 b may be measured bysetting the tri-state switches 911 a and 911 b to the second state, thetri-state switch 911 b to third state, and all other tri-state switchesof the plurality of tri-state switches 911 a, 911 b, . . . , 911 n and941 a, 941 b, . . . , 941 n to the first state. From a circuitcomplexity perspective, a tri-state switch may be equivalent to adding athird multiplexer circuit.

A further implementation variant of the circuit 1000 (not shown herein)omits the output multiplexer circuit 940 e.g., for reasons of complexityand cost reduction. In theory, the input multiplexer circuit 910 may beenough to selectively (e.g., sequentially) apply the voltage V₁ to asense circuit (e.g., sense circuit 106 a) and to selectively measure thecurrent I₂ at its measurement port 937 in response to the voltage V₁.Because the measurement amplifier circuit 404 is configured as a currentsummation amplifier, its output voltage V_(out) is indicative of the sumof the currents at each of the plurality of measurement ports 937. Asthe currents I₂ at the inactive sense circuits (e.g., sense circuit 106b) (where no voltage V₁ is applied) is ideally zero, the output voltageV_(out) is indicative of the current I₂ of the active sense circuit(e.g., sense circuit 106 a) where the voltage V₁ is applied. However, inpractice, disturbance currents e.g., capacitively coupling into thesense element (e.g., inductive sense element 107 b) when WPT is activemay also sum up resulting in a lower SNR as compared to a circuit 1000using the output multiplexer circuit 940.

A specification and some resulting performance figures of an examplecircuit 1000 with respect to the inductive sense circuits 106 configuredfor a nominal sense frequency ƒ_(s)=3 MHz are given in TABLE 9. A WPToperating frequency ƒ_(WPT)=85 kHz is assumed. TABLE 9 also includes thesystem Q-factor Q_(sys) as defined above by Equation (352), the qualityof the measurement circuit 104 defined as the ration:

Q _(mc) ≈|ΔV _(out) ′|/|ΔV ₁′|  (356)

|ΔV_(out)′| denotes the magnitude fractional change of the measurementamplifier circuit's 404 output voltage V_(out) caused by an object(e.g., object 110) and ΔV₁′ the fractional change of the drivercircuit's 402 output voltage V₁ caused by that object. Further, itincludes the degradation of the fractional change in the measurementamplifier circuit's 404 output voltage V_(out) as defined by Equation(354).

TABLE 9 Item Symbol Value Remarks Input multiplexer circuit 910 switchclosed- 5 Ω state resistance Parallel resistor 1014 resistance R_(par1)620 Ω Inductive sense element 107a inductance L 5 μH Inductive senseelement 107a equivalent R 3.8 Ω Q-factor Q = 25 series resistanceParallel capacitor 544 capacitance C_(p) 469 pF Series capacitor 546capacitance C_(s) 94 pF Capacitance ratio C_(p)/C_(s) n_(C) 5 Parallelresistor 1044 resistance R_(par2) 220 Ω Output multiplexer circuit 940switch 5 Ω closed-state resistance Feedback resistor 956 resistanceR_(f) 100 Ω Feedback capacitor 958 capacitance C_(f) 560 pF Sensecircuit 106a parallel resonance 3.29 MHz Precise sense frequency 2.9886MHz Transimpedance (minimum magnitude) Z₂₁₀ 130.8 Ω Meas. port 936 to937 System Q-factor Q_(sys) 21.6 Quality of measurement circuit 104Q_(mc) 27.3 Residual angle error ε −1.7° Degradation of fractionalchange at meas. 6.7% circuit 104 output (V_(out)) Driver amplifiercircuit 902 output voltage 2 V_(pp) 0.7 V_(rms) Driver amplifier circuit902 output current 6 mA_(rms) Driver amplifier circuit 902 output power4.2 mW Driver circuit 402 output current 5 mA_(rms) Inductive senseelement 107a current I_(L) 28.2 mA_(rms) (sense current) Transimpedanceamplifier circuit 952 input I_(in) 4.9 mA_(rms) 3 MHz sense signalcurrent Measurement amplifier circuit 404 output V_(out) 0.95 V_(pp) 3MHz sense signal, voltage 0.33 V_(rms) WPT fundamental disturbancevoltage V_(sW) 30 V_(rms) 85 kHz component induced into inductive senseelement 107a WPT fundamental disturbance voltage at 0.42 V_(pp) 85 kHzcomponent measurement amplifier circuit 404 output SNR with respect toWPT fundamental SNR_(W) 7.1 dB 85 kHz component disturbance voltage atf_(WPT) WPT fundamental disturbance voltage 4.6 V_(pp) 85 kHz componentacross input multiplexer circuit 910 switch when open WPT fundamentaldisturbance voltage 0.9 V_(pp) 85 kHz component across outputmultiplexer circuit 940 switch when open

A specification and some resulting performance figures of an examplecircuit 1000 with respect to the capacitive sense circuits 108configured for a nominal sense frequency ƒ_(s)=3 MHz are given in TABLE10. A WPT operating frequency ƒ_(WPT)=85 kHz is assumed. TABLE 10 alsoincludes the system Q-factor Q_(sys) as defined above by Equation (352),the quality of the measurement circuit 104 as defined by Equation (356),and the degradation of the fractional change in the measurementamplifier circuit's 404 output voltage V_(out) as defined by Equation(354).

TABLE 10 Item Symbol Value Remarks Input multiplexer circuit 910 switchclosed- 5 Ω state resistance Parallel resistor 1014 resistance R_(par3)1 kΩ DC block capacitor 930 capacitance C_(b) 2 nF Capacitive senseelement 109n capacitance C 30 pF Series inductor 724 inductance L_(s) 94uH includes transformer 726 leakage inductance Series inductor 724equivalent series R_(Ls) 118 Ω Includes transformer 726 resistancelosses, Q-factor Q_(Ls) ≈ 15 Transformer 726 secondary-referred mainL_(m) 150 uH inductance Transformer 726 voltage transformation 1:n_(VT)1:1 ratio Parallel resistor 1044 resistance R_(par4) 200 Ω Outputmultiplexer circuit 940 switch 5 Ω closed-state resistance Feedbackresistor 956 resistance R_(f) 100 Ω Feedback capacitor 958 capacitanceC_(f) 560 pF Sense circuit 106a parallel resonance 3 MHz Precise sensefrequency 3.0255 MHz Transimpedance (minimum magnitude) Z₂₁₀ 116 ΩSystem Q-factor Q_(sys) 14.2 Quality of measurement circuit 104 Q_(mc)25.2 Residual angle error ε 0.8° Degradation of fractional change atmeas. 4.1% circuit 104 output (V_(out)) Driver amplifier circuit 902output voltage 1.98 V_(pp) 0.7 V_(rms) Driver amplifier circuit 902output current 6.2 mA_(rms) Driver amplifier circuit 902 output power4.4 mW Capacitive sense element 109a current I_(C) 5.5 mA_(rms) (sensecurrent) Transimpedance amplifier circuit 952 input I_(in) 5.4 mA_(rms)3 MHz sense signal current Measurement amplifier circuit 404 outputV_(out) 1.05 V_(pp) 3 MHz sense signal, voltage 0.37 V_(rms) WPTfundamental disturbance voltage V_(sW) 150 V_(rms) 85 kHz componentinduced into capacitive sense element 109a WPT fundamental disturbancevoltage at 62 mV_(pp) 85 kHz component measurement amplifier circuit 404output SNR with respect to WPT fundamental SNR_(W) 24.6 dB 85 kHzcomponent disturbance voltage WPT fundamental disturbance voltage 2.3V_(pp) 85 kHz component across input multiplexer circuit 910 switch whenopen WPT fundamental disturbance voltage 0.07 V_(pp) 85 kHz componentacross output multiplexer circuit 940 switch when open

Comparing Tables 9 and 10 with Tables 11 and 12, respectively, shows thecircuit 1000 more potential if a higher sense current level I_(L) andI_(C) and higher power efficiency was targeted. While an amplifier 904output voltage constraint may limit the sense currents I_(L) and I_(C)in the circuit 900, an amplifier 954 input current constraint may limitthese currents in the circuit 1000.

A further aspect to be considered when comparing the circuits 900 and1000 are cross-coupling effects between neighboring inductive senseelements (e.g., inductive sense elements 107 a and 107 b of the array107). Cross-coupling may degrade the Q-factor of a sense circuit (e.g.,sense circuit 106 a or 108 n) due to energy absorption and may alsodistort its impedance function Z₁₁(ω) eventually compromisingperformance and the impedance angle measurement accuracy of themultipurpose detection circuit 100. This may be particularly true in acircuit (e.g., circuit 1000) using a plurality of sense circuits 106 or108 which, when inactive (e.g., deselected by the input and outputmultiplexer circuits 910 and 940, respectively), exhibit a parasiticparallel resonance as given by Equation (117) close to the sensefrequency. For the example implementation of the circuit 900 specifiedin TABLE 7, the parasitic parallel resonance may be at 1.3 MHz, whilefor the example implementation of the circuit 1000 specified in TABLE 9,it may occur close to the sense frequency at 3.29 MHz. Therefore, theimpact of cross-coupling in the circuit 1000 may be more significantthan in the circuit 900.

In an example implementation of the circuit 1000 using an array of sensecoils (e.g., array 107), a cross-coupling effect between inductive sensecircuits (e.g., sense circuit 106 a and 106 b) is reduced by configuringthe inductive sense circuits with a lower capacitance ration_(C)=C_(p)/C_(s) resulting in a lower impedance |Z₁₁| e.g., bytrading-off the impact of cross-coupling and thermal drift of the inputand output multiplexer circuits 910 and 940, respectively, as previouslydiscussed with reference to FIG. 10.

In another example implementation of the circuit 1000 using an array ofsense coils (e.g., array 107), a cross-coupling effect between inductivesense circuits (e.g., sense circuit 106 a and 106 b) is reduced byconfiguring (tuning) the inductive sense circuits associated withneighboring sense elements (e.g., sense element 107 a and 107 b) to adifferent resonant frequency e.g., following a frequency reuse scheme.However, this approach may cause a conflict in a multipurpose detectioncircuit 100 used to detect a passive beacon transponder (e.g., passivebeacon transponder 313 of FIG. 3) e.g., for purposes of detectingpresence of a vehicle (e.g., vehicle 330), a type of vehicle, or fordetermining a position of a vehicle. In some implementations, detectionof a passive beacon transponder requires each of the plurality ofinductive sense circuits 106 to be configured (tuned) to a substantiallyequal resonant frequency.

In a further example implementation using a planar array of sense coils(e.g., array 107), a gap is introduced between adjacent sense coils(e.g., between sense coil 107 a and 107 b) to reduce a cross-couplingeffect.

In a further aspect of an inductive sense coil array (e.g., array 107),loss effects in the lead lines are considered. In certainimplementations, the sense coil (e.g., sense coil 107 a) may beconnected to the capacitor 544 of the associated sense circuit (e.g.,sense circuit 106 a) via a long lead line. This may apply to animplementation of the circuit 1000 where the array of sense coils (e.g.,array 107) is carried on a separate printed circuit board (PCB)excluding any other components of the circuit 1000. A long lead line maycause substantial electrical losses degrading the Q-factor of a sensecircuit (e.g., sense circuit 106 a) and hence the performance of themultipurpose detection circuit 100.

In an example implementation of the circuit 1000 of FIG. 10 using asense coil array (e.g., array 107) e.g., implemented on a separate PCB,lead line losses are reduced by placing at least the parallel capacitor544 at a position close to the terminals of the sense coil (e.g., sensecoil 107 a). In such an implementation, the lead line of the sensecircuit 106 a may be between the series capacitor 546 and the parallelcircuit of capacitor 544 and the sense coil 107 a. Since the capacitors544 and 546 transform the impedance, the driver circuit 402 outputcurrent and thus the lead line current as required in the circuit 1000to generate a specified sense coil current I_(L) may be substantiallylower than the corresponding currents in the circuit 900. This is shownby numerical values for the driver circuit 402 output current as givenin TABLEs 7 and 9, respectively. Increasing the capacitance ration_(C)=C_(p)/C_(s), may reduce the lead line losses and hence increasingthe Q-factor of a sense circuit (e.g., sense circuit 106 a).

In an example implementation based on the circuit 900 of FIG. 9A using asense coil array (e.g., array 107) carried on a separate PCB, lead linelosses are reduced by placing the series capacitor 504 and an additionaltransformer (e.g., transformer 526 with reference to FIG. 5B) providinga transformation ratio n_(T):1 at a position close to the sense coil(e.g., sense coil 107 a). Increasing n_(T), may reduce the lead linelosses and hence increasing the Q-factor of a sense circuit (e.g., sensecircuit 106 a).

FIG. 11 is a circuit diagram of a circuit 1100 illustrating anotherexample implementation of a portion of a multi-purpose detection circuit100. The circuit 1100 of FIG. 11 illustrates an analog front-end circuitportion of the multi-purpose detection circuit 100 with reference toFIGS. 1 and 4. FIG. 11 excludes various other signal generation,processing, control and evaluation circuits (e.g., as shown in FIG. 4)that may be needed in some implementations of a multi-purpose detectioncircuit 100. The circuit 1100 implements inductive and capacitivesensing by measuring an impedance based on the current source voltagemeasurement approach as previously described in connection with FIG. 5C.

The circuit 1100 may be subdivided into an analog front-end part of themeasurement circuit 104 and a plurality of inductive and capacitivesense circuits 106 and 108 as previously described with reference to thegeneric block diagram of FIG. 4. The analog-front end part of themeasurement circuit 104 merges the driver circuit 402 and themeasurement amplifier circuit 404 with reference to FIGS. 1 and 4 anduses a single input multiplexer circuit 910 in common to selectively(e.g., sequentially) apply a voltage V₁ to each of the plurality ofsense circuits 106 and 108 and to measure a current I₁ in response tothe applied voltage V₁ as indicated in FIG. 11. The plurality ofinductive sense circuits 106 includes sense circuits 106 a, 106 b, . . ., 106 n (106 n not shown in FIG. 11 for purposes of illustration). Theplurality of capacitive sense circuits 108 includes sense circuits 108a, 108 b, . . . , 108 n, (108 a and 108 b not shown in FIG. 11 forpurposes of illustration). The dots indicated in FIG. 11 shall indicatethat the number of inductive sense circuits 106 and/or the number ofcapacitive sense circuits 108 may be greater than three as previouslynoted with reference to FIG. 1.

In the example implementation shown in FIG. 11, each of the plurality ofinductive sense circuits 106 have an identical circuit topology.Likewise, each of the plurality of capacitive sense circuits 108 have anidentical circuit topology. Therefore, descriptions given below for theinductive sense circuit 106 a also apply to the other inductive sensecircuits (e.g., 106 b) and descriptions given below for the capacitivesense circuit 108 n also apply to the other capacitive sense circuits(e.g., 108 a).

Each of the plurality of inductive sense circuits 106 provides ameasurement port 936 (indicated in FIG. 11 by a terminal) for applyingan electrical voltage V₁ (as indicated in FIG. 11) and for measuring anelectrical current I₁ (as indicated in FIG. 11) in response to thevoltage V_(I). Likewise, each of the plurality of capacitive sensecircuits 108 provides a measurement port 938 (indicated in FIG. 11 by aterminal) for applying the voltage V₁ (as indicated in FIG. 11) and formeasuring the current I₁ (as indicated in FIG. 11) in response to thecurrent V₁. The sense circuits 106 and 108 as illustrated in FIG. 11 maybe considered as one-port circuits.

As opposed to the circuits 900 and 1000, the measurement circuit 104 ofthe circuit 1100 includes a single input multiplexer circuit 910 toselectively (e.g., sequentially) apply the voltage V₁ to each of theplurality of sense circuits 106 and 108 and to selectively (e.g.,sequentially) measure a current I₁ in response to the applied voltageV₁. More specifically, but not indicated in FIG. 10 for purposes ofillustration, the voltage V₁ applied to the sense circuit 106 a may bedenoted by V_(1a), the voltage V₁ applied to the sense circuit 106 b maybe denoted as V_(1b), etc. Likewise, the currents I₁ in the sensecircuits 106 a and 106 b may be denoted by I_(1a) and I_(1b),respectively.

The circuit 1100 may be configured and operated in a mode to selectively(e.g., sequentially) measure the impedance Z₁₁ e.g., at the measurementport 936 of each of the plurality of the sense circuits 106 defined as:

Z ₁₁ ≈V _(1a) /I _(1a)  (357)

However, as opposed to the circuit 900 of FIG. 9A, the circuit 1100 asillustrated in FIG. 11 may be less versatile. It may not supportmeasuring the inter-sense circuit transimpedance Z₁₂ as described withreference to FIG. 9A.

The inductive sense circuit 106 a includes an inductive sense element107 a including a sense coil (e.g., sense coil 502 of FIG. 5A) with aninductance L, a first capacitor (e.g., capacitor 544 of FIG. 5C) havinga capacitance C_(p), and a second capacitor (e.g., capacitor 546 of FIG.5C) with a capacitance C. The first capacitor (e.g., parallel capacitor544) is electrically connected in parallel to the inductive senseelement 107 a. The parallel circuit of capacitor 544 and inductive senseelement 107 a electrically connects to ground and is also capacitivelycoupled to the measurement port 936 via the series capacitor 546. Asopposed to the sense circuits 106 of the circuit 900 of FIG. 9A, thesense circuits 106 of the circuit 1100 may not require an inductor(e.g., inductor 506) other than the inductive sense element (e.g.,inductive sense element 107 a) since the capacitor 546 may suffice toprovide the required high pass filter characteristic.

In an example implementation of the circuit 1100 of FIG. 11, each of theplurality of inductive sense circuits 106 is configured to provide aminimum of the impedance magnitude |Z₁₁| (series resonance)substantially at the sense frequency as previously discussed withreference to the circuit 540 of FIG. 5C.

In another aspect, the capacitors 544 and 546 may be configured toprovide an impedance magnitude |Z₁₁| at the sense frequency in asuitable range for the measurement circuit (e.g., measurement circuit104 of FIG. 4) by adjusting the capacitance ratio n_(C)=C_(p)/C_(s)accordingly as previously discussed with reference to FIG. 5G. Byincreasing the ratio n_(C), the impedance |Z₁₁| may be increased to alevel substantially (e.g., 10 times) higher than that of the inductivesense circuit 106 a of the circuit 900 of FIG. 9A. As previouslydiscussed with reference to FIG. 10, a higher impedance |Z₁₁| may be arequirement of the voltage source current measurement technique.

In some implementations, the capacitors 544 and 546 are of a type with alow temperature coefficient providing high thermal stability (e.g., aNP0-type capacitor) reducing thermal drift of an electricalcharacteristic (e.g., an impedance) as measured at each of the pluralityof inductive sense circuits 106 a, 106 b, . . . , 106 n.

Moreover, as previously discussed in connection with FIGS. 5C, 9A, and10, the inductive sense circuit 106 a in conjunction with the voltagesource current measurement technique provides a high pass filtercharacteristic to attenuate a low frequency disturbance component in thecurrent I₁. The series capacitor 546 together with the parallelcapacitor 544 may be configured to attenuate this low frequencydisturbance component to a level e.g., below the level of the current I₁in response to a respective sense voltage V₁ at the sense frequency,while maintaining series resonance at the sense frequency as previouslydiscussed with reference to FIG. 5G. At the sense frequency, this highpass filter characteristic may exert a minor impact on the current I₁and thus on the measurement of the impedance Z₁₁ and which may becorrected in a further processing (e.g., in the signal processingcircuit 408 of FIG. 4) as previously discussed with reference to thecircuit 900 of FIG. 9A.

In another aspect, the inductive sense circuits 106 may not need anysupplementary capacitor (e.g., capacitor 928) for purposes of DCblocking as previously discussed with reference to the circuit 900 ofFIG. 9A, as the series capacitor 546 already blocks any DC.

The capacitive sense circuit 108 n includes a capacitive sense element109 n including a sense electrode (e.g., sense electrode 702 of FIG. 7Cillustrating a single-ended sense electrode) with a capacitance C, aseries inductor 724 (e.g., series inductor 724 of FIG. 7C) with aninductance L a transformer 726 (e.g., transformer 726 of FIG. 7C)providing a primary and secondary port, a secondary-referred maininductance L_(m) and a voltage transformation ratio 1:n_(VT). Further,it includes a capacitor 930 with a capacitance C_(b) e.g., for purposesas previously discussed with reference to the sense circuits 106 and 108of FIG. 9A. As opposed to the sense circuit 108 n of FIG. 10, the sensecircuit 108 n of FIG. 11 provides only one measurement port 938 formeasuring the impedance Z₁₁. The inductor 724 is electrically connectedin series to the capacitive sense element 109 n. The series circuit ofinductor 724 and capacitive sense element 109 n is electricallyconnected to the transformer's 726 secondary port that also electricallyconnects to ground. Further, the transformer's 726 primary port iscapacitively coupled to the measurement port 938 via capacitor 930 andalso connects to ground.

In an example implementation of the circuit 1100 of FIG. 11, each of theplurality of capacitive sense circuits 108 is configured to provide aminimum of the impedance magnitude |Z₁₁| (series resonance)substantially at the sense frequency as previously discussed withreference to the circuit 720 of FIG. 7C.

In another aspect, the transformer 726 may be configured to provide animpedance magnitude |Z₁₁| at the sense frequency in a suitable range forthe measurement circuit (e.g., measurement circuit 104 of FIG. 4) byadjusting the transformation ratio 1:n_(VT) accordingly as previouslydescribed with reference to FIGS. 9A and 10.

The secondary-referred main inductance L_(m) of the transformer 726together with the capacitance C of the capacitive sense element 109 nform a 2^(nd) order high pass filter to attenuate a low frequencydisturbance component in the current I₁ as previously discussed withreference to FIG. 7C. The transformer 726 may be configured to attenuatethis low frequency disturbance component to a level e.g., significantlybelow the level of the current I₁ in response to a respective sensevoltage V₁ at the sense frequency. At the sense frequency, this highpass filter may exert a minor impact on the current I and thus on themeasurement of the impedance Z₁₁ and which may be corrected in a furtherprocessing (e.g., in the signal processing circuit 408 of FIG. 4) aspreviously discussed with reference to FIG. 7A.

The measurement circuit 104 includes a driver amplifier circuit 902, atransimpedance amplifier circuit (e.g., amplifier 904), a transformer1102, an input multiplexer circuit 910 illustrated in FIG. 11 as aplurality of switches 911 a, 911 b, . . . , 911 n, and a plurality ofresistors 1014 and 1015 (parallel resistors). The parallel resistors1014 and 1015 are electrically connected in parallel to the respectiveterminal of the input multiplexer circuit 910 whose terminals areelectrically connected to the respective measurement port (e.g.,measurement port 936) of the sense circuits 106 and 108. The parallelresistors 1014 connected in parallel to the measurement ports 936 have aresistance R_(par1), while the parallel resistors 1015 connected inparallel to the measurement ports 938 have a resistance R_(par2) thatmay generally differ from R_(par1).

Further, the measurement circuit 104 is configured to operate as avoltage source (e.g., voltage source 552 as described in connection withFIG. 5C) and to selectively (e.g., sequentially) apply the voltagesignal V₁ at the sense frequency to each of the plurality of inductivesense circuits 106 and to each of the plurality of the capacitive sensecircuits 108. The voltage signal V₁. (e.g., a sinusoidal sense signal)is based on a driver input signal which may be an output of the signalgenerator circuit 406 with reference to FIG. 4. The driver amplifiercircuit 902 as illustrated in FIG. 11 by example includes an amplifier904 and external resistance circuitry comprising a first (feedback)resistor 906 and a second resistor 908 for adjusting a gain. In someimplementations, the amplifier 904 is one of a low noise operationalamplifier and an amplifier providing high linearity. The driveramplifier circuit 902 is configured to receive the driver input signaland to provide a corresponding output with an accurate and stablevoltage (a voltage source output).

The measurement circuit 104 is also configured to operate as the analogfront-end part of a current measurement circuit (e.g., currentmeasurement circuit 550 as described in connection with FIG. 5C). It isconfigured to selectively (e.g., sequentially) buffer and convert anelectrical current I₁ in each of the plurality of inductive sensecircuits 106 and in each of the plurality of capacitive sense circuits108 and to provide an output voltage signal V_(out) (as indicated inFIG. 11) based on the respective electrical current I₁ at a levelsuitable for further processing e.g., in the signal processing circuit408 with reference to FIG. 4. The example transimpedance amplifiercircuit 952 as illustrated in FIG. 10 includes an amplifier 954, afeedback resistor 956 having a resistance R_(f), and a feedbackcapacitor 958 having a capacitance C_(f). In some implementations, theamplifier 954 is at least one of a low noise operational amplifier andan amplifier providing high linearity. The positive input (+) of theamplifier 954 connects to ground. Both the feedback resistor 958 and thefeedback capacitor 958 are electrically connected between the output(V_(out)) and the negative input (−) of the amplifier 954. Further, thetransimpedance amplifier circuit 952 is configured to receive an inputcurrent I_(in) that is a secondary current of the transformer 1102 andto convert the input current I_(in) into a proportional output voltageV_(out). The conversion gain (transimpedance) is determined by theimpedance of the parallel connection of the feedback resistor 956 andthe feedback capacitor 958. Since the voltage at the negative input (−)of the amplifier 954 is virtually zero (virtual ground), thetransimpedance amplifier circuit 952 presents a virtually zero inputimpedance at its negative input (−). The feedback capacitor 958 providesthe transimpedance amplifier circuit 952 with a first order low passfilter characteristic to attenuate disturbance signal components atfrequencies higher than the sense frequency (e.g., high order WPTharmonics) as previously discussed with reference to FIG. 9A.

In other implementations, the filtering of the transimpedance amplifiercircuit 952 is further enhanced by a supplementary feedback inductor(not shown in FIG. 10) electrically connected in parallel to thefeedback capacitor 958. In a further implementation, an active bandpassfilter as previously discussed with reference to FIG. 9A is employed.

The transformer 1102 includes a primary winding and a galvanicallyinsulated secondary winding wound on a common core as indicated in FIG.11. A first terminal of the transformer's 1102 primary windingelectrically connects to the output of the driver amplifier circuit 902,while its second terminal is electrically connected to a common node ofthe input multiplexer circuit 910. The transformer 1102 is configured asa current transformer providing a current transformation ratio 1:n_(CT).The transformer's primary current corresponding to the driver amplifiercircuit's 902 output current I may be indicative for the current I₁ inresponse to the applied voltage V₁. In some implementations, thetransformer 1102 is configured for n_(CT)>1 (e.g., in the range between2 and 4) and used to reduce a current I_(in)=I/n_(CT) at the input ofthe transimpedance amplifier circuit 952. Therefore, use of thetransformer 1102 with n_(CT)>1 may relax a drive current constrainsimilarly to the input transformer as discussed in connection with thecircuit 1000 of FIG. 10.

In some alternative implementations of the circuit 900 of FIG. 9A, thetransformer 1102 is supplanted by a parallel inductor (e.g., parallelinductor 716 with reference to the circuit 710 of FIG. 7B. In furtherimplementations, neither a transformer 1102 nor a parallel inductor isused.

The input multiplexer circuit 910 includes a plurality of switches 911a, 911 b, . . . , 911 n and is configured to selectively connect each ofthe plurality of inductive sense circuits 106 and each of the pluralityof capacitive sense circuits 108 to the driver amplifier circuit 902 andthe transimpedance amplifier circuit 952 to selectively (e.g.,sequentially) apply the voltage V₁ at the sense frequency to each of theplurality of sense circuits 106 and 108 and to selectively (e.g.,sequentially) measure the current I₁ in response to the applied voltageV₁. The input multiplexer circuit 910 is further configured to receive aMUX control signal from a control circuit (e.g., from the control andevaluation circuit 102 of FIG. 4) that controls the input multiplexercircuit 910.

Each of the plurality of switches 911 a, 911 b, . . . , 911 n may be oneof a type as previously mentioned with reference to the circuit 900 ofFIG. 9A. It may be appreciated that the closed-state resistance of theswitch (e.g., switch 911 a) may be non-critical if the impedancemagnitude |Z₁₁| at series resonance of a sense circuit (e.g., sensecircuit 106 a) is substantially higher than the closed-state resistance(e.g., if the capacitance ratio n_(C) is sufficiently large aspreviously discussed). A high enough impedance |Z₁₁| may also reduce animpact of the switch (e.g., switch 911 a) and thus improve a temperaturestability of the measurement circuit 104 as previously discussed withreference to FIG. 9A. If the impedance at the transformer's 1102 primaryport is small enough, the total capacitive load produced by theplurality of parallel capacitances at the common input/output node maybe non-critical given the voltage source output of the driver amplifiercircuit 902.

As in the circuit 1000 of FIG. 10, the parallel resistors 1014 and 1015are used to limit the open-switch voltage (e.g., at WPT frequency)across the input multiplexer circuit 910 switch (e.g., switch 911 a)when WPT is active as previously discussed with reference to FIG. 10.Decreasing the resistance (e.g., R_(par1)) of the parallel resistor(e.g., parallel resistor 1014) will reduce the open-switch voltage.However, it will also increase a voltage drop across the switch (e.g.,switch 911 a) when in closed-state impacting temperature stability ofthe measurement circuit 104 as previously discussed with reference toFIG. 9A. On the other hand and as opposed to the circuit 1000, there isonly one switch (e.g., switch 911 a) affecting temperature stability.Further, since the resistors (e.g., parallel resistor 1014) are inparallel to the respective measurement port (e.g., measurement port936), the impedance as measured by the measurement circuit 104 maydiffer from the impedance |Z₁₁| as presented e.g., at the measurementport 936. Moreover, the fractional change as measured in the measurementcircuit 104 may be smaller than the fractional change |ΔZ′| in |Z₁₁|.Therefore, in some implementations, the resistances R_(par1) andR_(par2) may represent a trade-off between an impedance measurementerror and an open-switch voltage. In some implementations, the inputmultiplexer circuit 910 switches (e.g., switch 911 a) are configured tosustain an open-switch voltage that is substantially (e.g., 5 times)higher than that of the switches used in the circuit 1000 and theresistances R_(par1) and R_(par2) are substantially (e.g., 5 times)higher than |Z₁₁|. Using an analog switch rated for a higher open-switchvoltage may require other electrical characteristics (e.g., closed-stateresistance, parasitic capacitances, etc.) of the switch to becompromised.

A specification and some resulting performance figures of an examplecircuit 1100 with respect to the inductive sense circuits 106 configuredfor a nominal sense frequency f_(s)=3 MHz are given in TABLE 11. A WPToperating frequency f_(WPT)=85 kHz is assumed. TABLE 10 also includesthe system Q-factor Q_(sys) as defined above by Equation (352), thequality of the measurement circuit 104 as defined by Equation (356), andthe degradation of the fractional change in the measurement amplifiercircuit's 404 output voltage V_(out) as defined by Equation (354).

TABLE 11 Item Symbol Value Remarks Input multiplexer circuit 910 switchclosed- 5 Ω state resistance Parallel resistor 1014 resistance R_(par1)620 Ω Inductive sense element 107a inductance L 5 μH Inductive senseelement 107a equivalent R 3.8 Ω Q-factor Q = 25 series resistanceParallel capacitor 544 capacitance C_(p) 450 pF Series capacitor 546capacitance C_(s) 113 pF Capacitance ratio C_(p)/C_(s) n_(C) 4Transformer 1102 current transformation 1:n_(CT) 1:3 ratio Feedbackresistor 956 resistance R_(f) 150 Ω Feedback capacitor 958 capacitanceC_(f) 330 pF Sense circuit 106a parallel resonance 3.35 MHz Precisesense frequency 2.991 MHz Impedance (min. magnitude) Z₁₁₀ 92 Ω Meas.port 936 System Q-factor Q_(sys)  19.7 Quality of measurement circuit104 Q_(mc) 16  Residual angle error ε  1° Degradation of fractionalchange at meas.   17.3% circuit 104 output (V_(out)) Driver amplifiercircuit 902 output voltage 2.23 V_(pp) 0.8 V_(rms) Driver amplifiercircuit 902 output current 9.4 mA_(rms) Driver amplifier circuit 902output power 7.5 mW Measurement circuit 104 output current I₁ 8.2mA_(rms) Inductive sense element 107a current I_(L) 39.6 mArms (sensecurrent) Transimpedance amplifier circuit 952 input I_(in) 3.1 mA_(rms)3 MHz sense signal current Measurement circuit 104 output voltageV_(out) 0.97 V_(pp) 3 MHz sense signal, 0.35 V_(rms) WPT fundamentaldisturbance voltage V_(sW) 30 V_(rms) 85 kHz component induced intoinductive sense element 107a WPT fundamental disturbance voltage at 0.25V_(pp) 85 kHz component measurement circuit 104 output SNR with respectto WPT fundamental SNR_(W) 11.7 dB 85 kHz component disturbance voltageat f_(WPT) WPT fundamental disturbance voltage 5.3 V_(pp) 85 kHzcomponent across input multiplexer circuit 910 switch when open

A specification and some resulting performance figures of an examplecircuit 1100 with respect to the capacitive sense circuits 108configured for a nominal sense frequency f_(s)=3 MHz are given in TABLE12. A WPT operating frequency f_(WPT)=85 kHz is assumed. TABLE 10 alsoincludes the system Q-factor Q_(sys) as defined above by Equation (352),the quality of the measurement circuit 104 as defined by Equation (356),and the degradation of the fractional change in the measurementamplifier circuit's 404 output voltage V_(out) as defined by Equation(354).

TABLE 12 Item Symbol Value Remarks Input multiplexer circuit 910 switchclosed- 5 Ω state resistance Parallel resistor 1015 resistance R_(par2)820 Ω DC block capacitor 930 capacitance C_(b) 3.3 nF Capacitive senseelement 109n capacitance C 30 pF Series inductor 724 inductance L_(s) 94uH includes transformer 726 leakage inductance Series inductor 724equiv. series resistance R_(Ls) 118 Ω Q-factor Q_(Ls) = 25, includestransf. 726 losses Transformer 726 secondary-referred main L_(m) 150 uHinductance Transformer 726 voltage transformation 1:n_(VT) 4:3 ratioTransformer 1102 current transformation 1:n_(CT) 1:3 ratio Feedbackresistor 956 resistance R_(f) 150 Ω Feedback capacitor 958 capacitanceC_(f) 330 pF Sense circuit 108n parallel resonance 1.86 MHz Precisesense frequency 3.009 MHz Impedance (min. magnitude) Z₁₁₀ 208 Ω Meas.port 938 System Q-factor Q_(sys) 11.6 Residual angle error ε  1°Degradation of fractional change at meas.  22.5% circuit 104 output(V_(out)) Quality of measurement circuit 104 Q_(mc) 33.2 Driveramplifier circuit 902 output voltage 4.24 V_(pp) 1.5 V_(rms) Driveramplifier circuit 902 output current I 8.8 mA_(rms) Driver amplifiercircuit 902 output power 13.2 mW Measurement circuit 104 output currentI₁ 11.5 mA_(rms) Capacitive sense element 109a current I_(C) 9.3mA_(rms) (sense current) Transimpedance amplifier circuit 952 inputI_(in) 2.9 mA_(rms) 3 MHz sense signal current Measurement circuit 104output voltage V_(out) 0.91 V_(pp) 3 MHz sense signal, 0.32 V_(rms) WPTfundamental disturbance voltage V_(sW) 150 V_(rms) 85 kHz componentinduced into capacitive sense element 109a WPT fundamental disturbancevoltage at 85.1 mV_(pp) 85 kHz component measurement circuit 104 outputSNR with respect to WPT fundamental SNR_(W) 20.5 dB 85 kHz componentdisturbance voltage at f_(WPT) WPT fundamental disturbance voltage 4.9V_(pp) 85 kHz component across input multiplexer circuit 910 switch whenopen

The numerical values as shown in TABLEs 11 and 12 for the circuit 1100are similar to those obtained for the circuit 1000 listed in TABLEs 9and 10, respectively. The degradation of the fractional change in thecircuit 1100 is larger. However, this drawback may be acceptableconsidering the potential for circuit complexity reduction in thecircuit 1100.

FIGS. 12A and 12B illustrate example implementations of capacitive senseelements 109 a, 109 b, . . . , 109 n integrated into the housing 328(e.g., a plastic enclosure) of a wireless power transfer structure(e.g., wireless power transfer structure 200) with reference to FIGS. 2and 3. FIGS. 12A and 12B show top views of the example implementations,which, for simplicity of discussion, include only a portion of thehousing 328.

More specifically, FIG. 12A shows an arrangement 1200 of foursingle-ended capacitive sense elements each composed of an electrodepair electrically connected in parallel and providing a single terminal1208 in the corner of the housing 328. Each electrode is furthersubdivided into smaller elements 1202 tailored to fit into compartments1204 of the housing 328. The plurality of elements 1202 belonging to thesame electrode are electrically connected e.g., using wires or similarelectrical conductors that may pass through slots in the walls dividingthe compartments 1204. The compartments 1204 are located along aperimeter of the housing 328. In some aspects, the top surface of thecompartments 1204 is inclined toward the interior of the housing 328 toform ramps over which a vehicle may drive.

FIG. 12B shows an arrangement 1210 of four double-ended capacitive senseelements each composed of an electrode pair providing a terminal pair1212 in the corner of the housing 328. Each electrode is furthersubdivided into smaller elements 1202 as described above with referenceto FIG. 12A.

FIGS. 13A and 13B illustrate example implementations of an electrodeelement (e.g., element 1202) with reference to FIGS. 12A and 12B,respectively. FIG. 13A illustrates holohedral sense electrode (e.g.,element 1202) made of a holohedral conductive sheet 1304 (e.g., coppersheet) on a non-conductive substrate 1302.

Integrated into a wireless power transfer structure (e.g., wirelesspower transfer structure 200), electrodes of capacitive sense elements109 a, 109 b, . . . , 109 n may experience substantial eddy currentheating due to the strong alternating magnetic fields as generatedduring wireless power transfer. Therefore, in a further aspect ofmitigating eddy current heating, electrodes are designed to increase asurface impedance.

FIG. 13B shows an alternative implementation of an element 1202configured to increase a surface impedance with respect to theholohedral conductive sheet 1204. The element 1202 is shaped as a fingerstructure comprising a number of conductive strips electricallyconnected at only one end. This finger structure may increase a surfaceimpedance and thus reduce eddy current flow on the element's 1202surface and consequent heating. Using a finger structure may also reducethe capacitance C of the capacitive sense element (e.g., capacitivesense element 109 a). However, the capacitive coupling to the object(e.g., object 114) e.g., as defined by Equation (342), may not changesubstantially.

In another implementation, the element 1202 is made of a weaklyconductive material providing a sufficiently high surface impedance. Thematerial may represent a trade-off between eddy current heating and anequivalent resistance (e.g., equivalent resistance R as indicated inFIG. 7A) of the capacitive sense element (e.g., capacitive sense element109 a).

In further implementations, other suitable structures or materials areused to increase the surface impedance of the element 1202 trading-offeddy current heating and an equivalent resistance R of the capacitivesense element (e.g., capacitive sense element 109 a) at the sensefrequency as previously discussed with reference to FIG. 7A.

In some implementations, the element 1202 is made as a printed circuitboard (PCB). In other implementations, the plurality of elements 1202 isa flex print that also includes the inter-element connections asmentioned above with reference to FIG. 12A.

In further implementations with reference to FIG. 12, the electrodes ofthe capacitive sense elements (e.g., capacitive sense element 109 a) aredirectly printed onto the non-conductive inner surface of the housing328 e.g., using a 3D inkjet printer (conductive ink) or othermanufacturing technologies such as Molded Interconnect Device (MID) andLaser Direct Structuring (LDS). In another implementation, electrodesare fully or partially embedded in the plastic material of the housing328 e.g., using an injection molding process where electrodes are inlaidinto the mold prior injection.

The image (snapshot) sequence of FIGS. 14A to 14C illustrate theelectric vehicle 330 driving into a parking place 1404 (e.g., forpurposes of charging) at three different positions. The vehicle 330provides the (secondary) wireless power transfer structure 310 mountedat its underbody at a location as illustrated in FIGS. 14A to 14C and inFIG. 3. The parking place 1404 is equipped with the (primary) wirelesspower transfer structure 200 with reference to FIGS. 2 and 3 e.g., towirelessly deliver power to the vehicle 330 as previously described withreference to FIGS. 2 and 3. Further, the wireless power transferstructure 200 integrates the multi-purpose detection circuit 100including the array 107 of inductive sense elements 107 a, 107 b, . . .107 n and the arrangement of capacitive sense elements 109 a, 109 b, . .. 109 n as illustrated in FIGS. 2 and 3. The array 107 of inductiveelements is assumed to essentially cover the top surface of the wirelesspower transfer structure 200 as illustrated by FIGS. 2 and 3.

FIG. 14A shows the vehicle 330 approaching the ground-based wirelesspower transfer structure 200. In FIG. 14B, the vehicle's 330 frontpartially overlaps the wireless power transfer structure 200, while FIG.14C represents the vehicle 330 close before its final parking positionwhere the vehicle-mounted wireless power transfer structure 310 will bein sufficient alignment with the ground-based wireless power transferstructure 200 as previously mentioned in the introduction.

The sequence of images 1500 to 1580 of FIG. 15A and 15B illustrate theelectric vehicle 330 advancing towards the final parking position asdescribed above with reference to FIG. 14C. On the right-hand side,FIGS. 15A and 15B display corresponding 8×8 pixel grayscale patterns1502 to 1592. Each of the patterns 1502 to 1592 may refer to a patternproduced by mapping the plurality of detection output values of themulti-purpose object detection circuit 100 using an 8×8 array 107 ofinductive sense elements 107 a, 107 b, . . . , 107 n onto respectiveelements of an 8×8 matrix at the respective vehicle position. Morespecifically, in some implementations, these detection outputs may referto outputs of the measurement circuit 104 with reference to FIG. 4. Inother implementations, detection outputs may be outputs of a function(not shown herein) that is part of the evaluation & control circuit 102.The pixel grayscale may be indicative of at least one of a magnitude anda phase of an impedance change (e.g., ΔZ) in an inductive sense circuit(e.g., inductive sense circuit 106 a) as caused by the presence of thevehicle 330. In some implementations, it may be indicative of anotherelectrical characteristics as output by the multi-purpose objectdetection circuit 100. “Dark gray” indicates zero change (e.g., |ΔZ|≈0)or a change blow a detection threshold, while “white” refers to a changereaching or exceeding a certain saturation level.

Pattern 1502 (all pixels dark gray) refers to the absence of the vehicle330 or to a vehicle 330 position as shown by image 1500 where thevehicle's 330 impact on the detection outputs of the multi-purposeobject detection circuit 100 is below the detection threshold.

Pattern 1512 refers to a vehicle 330 position as shown by image 1510where the front (leading edge) of the vehicle 330 starts to cause aminority of detection outputs to exceed the detection thresholdresulting in brighter gray pixels in the 1^(st) and 2^(nd) column of thepattern 1512.

Pattern 1522 refers to a vehicle 330 position as shown by image 1520where the leading edge of the vehicle 330 causes more detection outputsto exceed the detection threshold or even the saturation level resultingin white pixels in the 1^(st) and 2^(nd) column and brighter gray pixelsin the 3^(rd) and 4^(th) column of the pattern 1522.

Pattern 1532 refers to a vehicle 330 position as shown by image 1530where the leading edge of the vehicle 330 is further advanced andsubstantially overlapping the surface of the wireless power transferstructure 200, thus causing a majority of detection outputs to exceedthe detection threshold and a higher number thereof to exceed thesaturation level resulting in white pixels in the first four columns andbrighter gray pixels in the 5^(th) and 6^(th) column of the pattern1532.

Pattern 1542 refers to a vehicle 330 position as shown by image 1540where the leading edge of the vehicle 330 is almost fully overlappingthe wireless power transfer structure 200, thus causing all detectionoutputs to exceed the detection threshold and a majority thereof toexceed the saturation level resulting in white pixels in the first 6columns and brighter gray pixels in the 7^(th) and 8^(th) column of thepattern 1542. At this stage, the pattern 1542 also shows a gray area inthe first two columns caused by an inhomogeneous structure of thevehicle's 330 underbody (e.g., by a different material or a cavity inthe underbody).

Pattern 1552 refers to a vehicle 330 position as shown by image 1550where the leading edge of the vehicle 330 entirely overlaps the wirelesspower transfer structure 200, thus causing detection outputs to exceedthe detection threshold in all columns. The gray area caused by theinhomogeneous underbody and that has become visible in the pattern 1542has now moved to the 4^(th) and 6^(th) column of the pattern 1552.

Pattern 1562 refers to a vehicle 330 position as shown by image 1560where the vehicle-based wireless power transfer structure 310 hasreached the edge of the ground-based wireless power transfer structure200 that starts now to also impact the pattern 1562. Since the wirelesspower transfer structure 310 includes different materials (e.g., Litzwire made of copper, ferrite, aluminum, and other conductive andnon-conductive materials, its impact on the individual inductive senseelements of the array 107 may be highly variable. While ferritematerials tend to produce a positive reactance change, highly conductivematerials such as copper and aluminum tend to cause a negative reactancechange. Depending on the actual relative position of the wireless powertransfer structure 310, the impact of some portions of the wirelesspower transfer structure 310 on some inductive sense elements (e.g.,inductive sense element 107 a) may cancel out producing the dark grayarea in column 1 of the pattern 1562.

Pattern 1572 refers to a vehicle 330 position as shown by image 1570where the center of the vehicle-based wireless power transfer structure310 has just surpassed the edge of the ground-based wireless powertransfer structure 200 producing a unique pattern of different graylevels in the first three columns. The gray area caused by the underbodyinhomogeneity has now proceeded to the last two columns of the pattern1572.

Pattern 1582 refers to a vehicle 330 position as shown by image 1580where the vehicle-based wireless power transfer structure 310 now fullyoverlaps with the top surface of the ground-based wireless powertransfer structure 200. The grayscale pattern as produced by thevehicle-based wireless power transfer structure 310 is now almostentirely visible, while the underbody inhomogeneity has already left thesensitive area of the inductive sensing array 107 (not visible anymorein the pattern of image 1580).

Pattern 1592 refers to a vehicle 330 position as shown by image 1590where the vehicle-based wireless power transfer structure 310 is nowwell aligned with the ground-based wireless power transfer structure 200displaying the grayscale pattern centered in the 8×8 pattern 1592.

The patterns 1502 to 1592 as used in FIGS. 15A and 15B may be consideredexample and simplified for purposes of illustration. Other types ofvehicles 330 and other types of vehicle-based wireless power transferstructures 310 may produce 8×8 patterns different from those shown inFIGS. 15A and 15B.

Patterns (e.g., 2×2 patterns) may also be produced from detectionoutputs of the multi-purpose detection circuit 100 associated to theplurality of capacitive sense elements 109 a, 109 b, . . . 109 n asillustrated in FIGS. 2 and 3. More specifically, in someimplementations, these detection outputs may refer to outputs of themeasurement circuit 104 with reference to FIG. 4. In otherimplementations, detection outputs may be outputs of a function (notshown herein) that is part of the evaluation & control circuit 102.Analogously, gray levels in these patterns (not shown herein) may beindicative of impedance changes as measured in the plurality ofcapacitive sense circuits 108 a, 108 b, . . . 108 n. Though with a lowerimage resolution, these patterns may also reflect structures of thevehicle's 330 underbody (e.g., the vehicle-based wireless power transferstructure 310).

Therefore, in some implementations, the patterns 1502 to 1592 may alsorefer to a pattern produced by mapping detection output values of themulti-purpose object detection circuit 100 associated with at least oneof the plurality of inductive sense elements 107 a, 107 b, . . , 107 nand the plurality of capacitive sense elements 109 a, 109 b, . . . , 109c.

In an aspect of the multi-purpose detection circuit 100, patterns asproduced by detection outputs associated with at least one of theplurality of inductive sense circuits 106 and the plurality ofcapacitive sense circuits as previously described with reference toFIGS. 15A and 15B are used to discriminate the vehicle 330 from anobject (e.g., object 110). More specifically, in some implementations,detection outputs associated with at least one inductive sense circuit(e.g., inductive sense circuit 106 a) are used to discriminate betweenthe impact of the vehicle 330 and the impact of an object (e.g., object112) in detection outputs associated with at least one capacitive sensecircuit (e.g., capacitive sense circuit 108 a). Conversely, in someimplementations, detection outputs associated with at least onecapacitive sense circuit (e.g., capacitive sense circuit 108 a) are usedto discriminate between the impact of the vehicle 330 and the impact ofan object (e.g., object 110) in detection outputs associated with atleast one inductive sense circuit (e.g., inductive sense circuit 106 a).

Stated more generally, detection outputs associated with at least oneinductive sense circuit (e.g., inductive sense circuit 106a) are used toreduce a false positive detection probability of LOD. Conversely,detection outputs associated with at least one capacitive sense circuit(e.g., capacitive sense circuit 108 a) are used to reduce a falsepositive detection probability of FOD.

In some implementations, detection outputs associated with at least oneinductive sense circuit (e.g., inductive sense circuit 106 a) and atleast one capacitive sense circuit (e.g., capacitive sense circuit 108a) are used to dynamically adjust a detection threshold of themulti-purpose detection circuit 100 where the detection threshold refersto at least one of FOD and LOD. Dynamically adjusting a detectionthreshold is described in U.S. patent application Ser. No. 16/392,464,titled Extended Foreign Object Detection Signal Processing.

In another aspect of the multi-purpose detection circuit 100, patternsas produced by detection outputs associated with at least one of theplurality of inductive sense circuits 106 and the plurality ofcapacitive sense circuits as previously described with reference toFIGS. 15A and 15B are used to detect or identify a type of vehicle 330or a type of vehicle-based wireless power transfer structures 310.

In a further aspect of the multi-purpose detection circuit 100, patternsas produced by detection outputs associated with at least one of theplurality of inductive sense circuits 106 and the plurality ofcapacitive sense circuits as previously described with reference toFIGS. 15A and 15B are used to determine a position of the vehicle 330(or the vehicle-based wireless power transfer structures 310) relativeto the ground-based wireless power transfer structure 200.

In some implementations of the multi-purpose detection circuit 100, therelative position is at least in part determined by using an imagecorrelation technique (e.g., a similar technique as employed in thecomputer mouse using a laser sensor for surface structure detection).

In another implementation of the multi-purpose detection circuit 100,the relative position is determined by tracking a “front-wave” insuccessively obtained patterns as produced by detection outputsassociated with at least one of the plurality of inductive sensecircuits 106 and the plurality of capacitive sense circuits and asillustrated by the patterns 1512 to 1542 of FIGS. 15A. This “front-wave”visible in patterns 1512 to 1542 as a transition of dark grey to whitemay be produced by the leading edge of the vehicle 330.

In another aspect of the multi-purpose detection circuit 100, patternsas produced by detection outputs associated with at least one of theplurality of inductive sense circuits 106 and the plurality ofcapacitive sense circuits as previously described with reference toFIGS. 15A and 15B are used to activate or prime another positioningsystem.

In a further aspect of the multi-purpose detection circuit 100, patternsas produced by detection outputs associated with at least one of theplurality of inductive sense circuits 106 and the plurality ofcapacitive sense circuits as previously described with reference toFIGS. 15A and 15B are used to extend the range of another positioningsystem.

In yet another aspect of the multi-purpose detection circuit 100,patterns as produced by detection outputs associated with at least oneof the plurality of inductive sense circuits 106 and the plurality ofcapacitive sense circuits as previously described with reference toFIGS. 15A and 15B are used to enhance the accuracy or reliability ofanother positioning system.

In yet a further aspect of the multi-purpose detection circuit 100,patterns as produced by detection outputs associated with at least oneof the plurality of inductive sense circuits 106 and the plurality ofcapacitive sense circuits as previously described with reference toFIGS. 15A and 15B are used to adjust or calibrate another positioningsystem.

Other positioning systems may include systems based on using at leastone of an inductive and capacitive passive beacon transponder aspreviously discussed e.g., with reference to FIG. 3, sensing of at leastone of a magnetic and electric field generated by an active beacontransmitter (e.g., active beacon as described in U.S. patent applicationSer. No. 16/284,959 titled Extended-Range Positioning System Based onForeign Object Detection, magnetic vectoring as described in U.S. patentapplication Ser. No. 15/003,521 titled Integration of SolenoidPositioning Antennas in Wireless Inductive Charging Power Applications,U.S. Pat. No. 10,340,752 titled System, Methods and Apparatuses forGuidance and Alignment in Electric Vehicles Wireless Inductive ChargingSystems, and U.S. Pat. No. 10,566,839 titled Systems, Methods andApparatus for Guidance and Alignment Between Electric Vehicles andWireless Charging Systems, the entire contents of which are herebyincorporated by reference). They may also include positioning systemsbased on optical sensors (cameras), LIDAR technologies, ultrasoundsensors, inertial sensors.

The various operations of methods described above may be performed byany suitable means capable of performing the corresponding functions.The means may include various hardware and/or software component(s)and/or module(s), including, but not limited to a circuit, anapplication-specific integrated circuit (ASIC), or processor. Generally,where there are operations illustrated in figures, those operations mayhave corresponding counterpart means-plus-function components withsimilar numbering.

As used herein, the term “determining” encompasses a wide variety ofactions. For example, “determining” may include calculating, computing,processing, deriving, investigating, looking up (e.g., looking up in atable, a database, or another data structure), ascertaining, and thelike. Also, “determining” may include receiving (e.g., receivinginformation), accessing (e.g., accessing data in a memory), and thelike. Also, “determining” may include resolving, selecting, choosing,establishing, and the like.

As used herein, a phrase referring to “at least one of” a list of itemsrefers to any combination of those items, including single members. Asan example, “at least one of: a, b, or c” is intended to cover: a, b, c,a-b, a-c, b-c, and a-b-c, as well as any combination with multiples ofthe same element (e.g., a-a, a-a-a, a-a-b, a-a-c, a-b-b, a-c-c, b-b,b-b-b, b-b-c, c-c, and c-c-c or any other ordering of a, b, and c).

The various illustrative logical blocks, modules and circuits describedin connection with the present disclosure may be implemented orperformed with a general-purpose processor, a digital signal processor(DSP), an ASIC, a field programmable gate array (FPGA) or otherprogrammable logic device (PLD), discrete gate or transistor logic,discrete hardware components, or any combination thereof designed toperform the functions described herein. A processor may be amicroprocessor, but in the alternative, the processor may be anycommercially available processor, controller, microcontroller, or statemachine. A processor may also be implemented as a combination ofcomputing devices, e.g., a combination of a DSP and a microprocessor, aplurality of microprocessors, one or more microprocessors in conjunctionwith a DSP core, or any other such configuration.

The methods disclosed herein comprise one or more steps or actions forachieving the described method. The method steps and/or actions may beinterchanged with one another without departing from the scope of theclaims. In other words, unless a specific order of steps or actions isspecified, the order and/or use of specific steps and/or actions may bemodified without departing from the scope of the claims.

The functions described may be implemented in hardware, software,firmware, or any combination thereof. If implemented in hardware, anexample hardware configuration may comprise a processing system in awireless node. The processing system may be implemented with a busarchitecture. The bus may include any number of interconnecting busesand bridges depending on the specific application of the processingsystem and the overall design constraints. The bus may link togethervarious circuits including a processor, machine-readable media, and abus interface. The bus interface may be used to connect a networkadapter, among other things, to the processing system via the bus. Thebus may also link various other circuits such as timing sources,peripherals, voltage regulators, power management circuits, and thelike.

It is to be understood that the claims are not limited to the preciseconfiguration and components illustrated above. Various modifications,changes and variations may be made in the arrangement, operation anddetails of the methods and apparatus described above without departingfrom the scope of the claims.

What is claimed is:
 1. An apparatus configured for foreign objectdetection, living object detection, vehicle detection, vehicle typedetection, and vehicle position detection, the apparatus comprising: amulti-purpose detection circuit having: a plurality of inductive sensecircuits forming an array; and a plurality of capacitive sense circuitssurrounding the array.
 2. The apparatus of claim 1, wherein each of theplurality of inductive sense circuits includes at least one inductivesense element and an associated capacitive element to compensate forgross reactance as presented at a plurality of terminals of the at leastone inductive sense element at a sense frequency.
 3. The apparatus ofclaim 2, wherein the at least one inductive sense element comprises asense coil.
 4. The apparatus of claim 1, wherein each of the pluralityof capacitive sense circuits includes at least one capacitive senseelement and an associated inductive element to compensate for grossreactance as presented at a plurality of terminals of the at least onecapacitive sense element at a sense frequency.
 5. The apparatus of claim4, wherein the at least one capacitive sense element comprises a senseelectrode.
 6. The apparatus of claim 1, wherein at least one sensecircuit of the plurality of inductive sense circuits and the pluralityof capacitive sense circuits includes an impedance matching element fortransforming an impedance of the at least one sense circuit to matchwith an operating impedance range of the apparatus.
 7. The apparatus ofclaim 6, wherein the impedance matching element comprises a transformer.8. The apparatus of claim 1, further comprising: a measurement circuitfor selectively and sequentially measuring an electrical characteristicin each of the plurality of inductive sense circuits and each of theplurality of capacitive sense circuits according to a predetermined timemultiplexing scheme.
 9. The apparatus of claim 8, wherein the electricalcharacteristic includes an impedance.
 10. The apparatus of claim 1,wherein: the plurality of inductive sense circuits are oriented in aplanar array integrated into a ground-based wireless power transferstructure; the plurality of capacitive sense circuits are positioned 11.The apparatus of claim 10, wherein: the housing includes a plurality ofcompartments located along a perimeter of the housing; and the pluralityof capacitive sense circuits each include a capacitive sense element;each capacitive sense element is subdivided into smaller elementstailored to fit into respective compartments of the plurality ofcompartments of the housing.
 12. The apparatus of claim 11, wherein oneor more of the smaller elements have a finger structure to reduce eddycurrent heating.
 13. The apparatus of claim 11, wherein the capacitivesense elements are electrically connected to one another in parallel andprovide one or more terminals in one or more corners of the housing. 14.The apparatus of claim 1, wherein the multi-purpose detection circuit isconfigured to activate a passive beacon detection based on a combinationof detections from a passive beacon on a vehicle and detections ofpatterns of impedance changes in the array that are associated withvehicle detection.
 15. A detection circuit comprising: a plurality ofcapacitive sense elements configured for living object detection; ameasurement circuit electrically connected to the plurality ofcapacitive sense elements, the measurement circuit configured to measurereflected impedance of an object in proximity to one or more of theplurality of capacitive sense elements, the reflected impedance measuredat a sense frequency of the plurality of capacitive sense elements; anda control and evaluation circuit electrically connected to themeasurement circuit, the control and evaluation circuit configured todiscriminate drops of water from living objects and one or morenon-living dielectric objects based on the reflected impedance toprevent a false positive detection due to the drops of water.
 16. Thedetection circuit of claim 15, wherein the drops of water arediscriminated from the living objects and the one or more non-livingdielectric objects based on an angle, corresponding to the reflectedimpedance of the drops of water in a complex plane, being within a firstangle range that is outside of a second angle range used to detect theliving objects and the one or more non-living dielectric objects. 17.The detection circuit of claim 16, wherein the angle is less than 60degrees.
 18. The detection circuit of claim 16, wherein the angle isassociated with a volume-to-surface area ratio of the drops of water.19. The detection circuit of claim 15, wherein: the detection circuit isintegrated with a wireless-power transfer structure; and the drops ofwater include one of rain or melt water dripping from a vehiclepositioned over the wireless-power transfer structure.
 20. The detectioncircuit of claim 15, wherein the control and evaluation circuit isconfigured to suppress detections caused by an impedance change, in theone or more of the capacitive sense elements, corresponding to a compleximpedance change that substantially deviates from 90 degrees.